LM13700M - NATIONAL SEMICONDUCTOR

LM13700

LM13700 Dual Operational Transconductance Amplifiers with Linearizing

Diodes and Buffers

Literature Number: SNOSBW2D

LM13700

Dual Operational Transconductance Amplifiers with

Linearizing Diodes and Buffers

General Description

The LM13700 series consists of two current controlled

transconductance amplifiers, each with differential inputs

and a push-pull output. The two amplifiers share common

supplies but otherwise operate independently. Linearizing

diodes are provided at the inputs to reduce distortion and

allow higher input levels. The result is a 10 dB signal-to-

noise improvement referenced to 0.5 percent THD. High

impedance buffers are provided which are especially de-

signed to complement the dynamic range of the amplifiers.

The output buffers of the LM13700 differ from those of the

LM13600 in that their input bias currents (and hence their

output DC levels) are independent of I

ABC

. This may result in

performance superior to that of the LM13600 in audio appli-

cations.

Features

adjustable over 6 decades

nExcellent g

linearity

nExcellent matching between amplifiers

nLinearizing diodes

nHigh impedance buffers

nHigh output signal-to-noise ratio

Applications

nCurrent-controlled amplifiers

nCurrent-controlled impedances

nCurrent-controlled filters

nCurrent-controlled oscillators

nMultiplexers

nTimers

nSample-and-hold circuits

Connection Diagram

Dual-In-Line and Small Outline Packages

00798102

Top View

Order Number LM13700M, LM13700MX or LM13700N

See NS Package Number M16A or N16A

June 2004

LM13700 Dual Operational Transconductance Amplifiers with Linearizing Diodes and Buffers

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage

LM13700 36 V

or ±18V

Power Dissipation (Note 2) T

= 25˚C

LM13700N 570 mW

Differential Input Voltage ±5V

Diode Bias Current (I

)2mA

Amplifier Bias Current (I

ABC

)2mA

Output Short Circuit Duration Continuous

Buffer Output Current (Note 3) 20 mA

Operating Temperature Range

LM13700N 0˚C to +70˚C

DC Input Voltage +V

to −V

Storage Temperature Range −65˚C to +150˚C

Soldering Information

Dual-In-Line Package

Soldering (10 sec.) 260˚C

Small Outline Package

Vapor Phase (60 sec.) 215˚C

Infrared (15 sec.) 220˚C

Electrical Characteristics (Note 4)

Parameter Conditions LM13700 Units

Min Typ Max

Input Offset Voltage (V

) Over Specified Temperature Range 0.4 4 mV

ABC

= 5 µA 0.3 4

Including Diodes Diode Bias Current (I

) = 500 µA 0.5 5 mV

Input Offset Change 5 µA ≤I

ABC

≤500 µA 0.1 3 mV

Input Offset Current 0.1 0.6 µA

Input Bias Current Over Specified Temperature Range 0.4 5 µA

Forward 6700 9600 13000 µmho

Transconductance (g

) Over Specified Temperature Range 5400

Tracking 0.3 dB

Peak Output Current R

=0,I

ABC

=5µA 5

=0,I

ABC

= 500 µA 350 500 650 µA

= 0, Over Specified Temp Range 300

Peak Output Voltage

Positive R

=∞,5µA≤I

ABC

≤500 µA +12 +14.2 V

Negative R

=∞,5µA≤I

ABC

≤500 µA −12 −14.4 V

Supply Current I

ABC

= 500 µA, Both Channels 2.6 mA

Sensitivity

Positive ∆V

/∆V

20 150 µV/V

Negative ∆V

/∆V

−

20 150 µV/V

CMRR 80 110 dB

Common Mode Range ±12 ±13.5 V

Crosstalk Referred to Input (Note 5) 100 dB

20 Hz <f<20 kHz

Differential Input Current I

ABC

= 0, Input = ±4V 0.02 100 nA

Leakage Current I

ABC

= 0 (Refer to Test Circuit) 0.2 100 nA

Input Resistance 10 26 kΩ

Open Loop Bandwidth 2 MHz

Slew Rate Unity Gain Compensated 50 V/µs

Buffer Input Current (Note 5) 0.5 2 µA

Peak Buffer Output Voltage (Note 5) 10 V

Note 1: “Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits.

Note 2: For operation at ambient temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance,

junction to ambient, as follows: LM13700N, 90˚C/W; LM13700M, 110˚C/W.

Note 3: Buffer output current should be limited so as to not exceed package dissipation.

LM13700

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Electrical Characteristics (Note 4) (Continued)

Note 4: These specifications apply for VS=±15V, TA= 25˚C, amplifier bias current (IABC) = 500 µA, pins 2 and 15 open unless otherwise specified. The inputs to

the buffers are grounded and outputs are open.

Note 5: These specifications apply for VS=±15V, IABC = 500 µA, ROUT =5kΩconnected from the buffer output to −VSand the input of the buffer is connected

to the transconductance amplifier output.

Schematic Diagram

One Operational Transconductance Amplifier

00798101

Typical Application

00798118

Voltage Controlled Low-Pass Filter

LM13700

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Typical Performance Characteristics

Input Offset Voltage Input Offset Current

00798138

00798139

Input Bias Current Peak Output Current

00798140 00798141

Peak Output Voltage and

Common Mode Range Leakage Current

00798142 00798143

LM13700

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Typical Performance Characteristics (Continued)

Input Leakage Transconductance

00798144 00798145

Input Resistance

Amplifier Bias Voltage vs.

Amplifier Bias Current

00798146 00798147

Input and Output Capacitance Output Resistance

00798148 00798149

LM13700

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Typical Performance Characteristics (Continued)

Distortion vs. Differential

Input Voltage

Voltage vs. Amplifier

Bias Current

00798150

00798151

Output Noise vs. Frequency

00798152

Unity Gain Follower

00798105

LM13700

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Typical Performance Characteristics (Continued)

Leakage Current Test Circuit Differential Input Current Test Circuit

00798106

00798107

Circuit Description

The differential transistor pair Q

and Q

form a transcon-

ductance stage in that the ratio of their collector currents is

defined by the differential input voltage according to the

transfer function:

(1)

where V

is the differential input voltage, kT/q is approxi-

mately 26 mV at 25˚C and I

and I

are the collector currents

of transistors Q

and Q

respectively. With the exception of

and Q

, all transistors and diodes are identical in size.

Transistors Q

and Q

with Diode D

form a current mirror

which forces the sum of currents I

and I

to equal I

ABC

(2)

where I

ABC

is the amplifier bias current applied to the gain

pin.

For small differential input voltages the ratio of I

and I

approaches unity and the Taylor series of the In function can

be approximated as:

(3)

(4)

Collector currents I

and I

are not very useful by themselves

and it is necessary to subtract one current from the other.

The remaining transistors and diodes form three current

mirrors that produce an output current equal to I

minus I

thus:

(5)

The term in brackets is then the transconductance of the

amplifier and is proportional to I

ABC

Linearizing Diodes

For differential voltages greater than a few millivolts, Equa-

tion (3) becomes less valid and the transconductance be-

comes increasingly nonlinear. Figure 1 demonstrates how

the internal diodes can linearize the transfer function of the

amplifier. For convenience assume the diodes are biased

with current sources and the input signal is in the form of

current I

. Since the sum of I

and I

is I

ABC

and the differ-

ence is I

OUT

, currents I

and I

can be written as follows:

Since the diodes and the input transistors have identical

geometries and are subject to similar voltages and tempera-

tures, the following is true:

(6)

Notice that in deriving Equation (6) no approximations have

been made and there are no temperature-dependent terms.

The limitations are that the signal current not exceed I

and that the diodes be biased with currents. In practice,

replacing the current sources with resistors will generate

insignificant errors.

Applications

Voltage Controlled Amplifiers

Figure 2 shows how the linearizing diodes can be used in a

voltage-controlled amplifier. To understand the input biasing,

it is best to consider the 13 kΩresistor as a current source

and use a Thevenin equivalent circuit as shown in Figure 3.

This circuit is similar to Figure 1 and operates the same. The

potentiometer in Figure 2 is adjusted to minimize the effects

of the control signal at the output.

LM13700

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Applications

Voltage Controlled Amplifiers (Continued)

For optimum signal-to-noise performance, I

ABC

should be as

large as possible as shown by the Output Voltage vs. Ampli-

fier Bias Current graph. Larger amplitudes of input signal

also improve the S/N ratio. The linearizing diodes help here

by allowing larger input signals for the same output distortion

as shown by the Distortion vs. Differential Input Voltage

graph. S/N may be optimized by adjusting the magnitude of

the input signal via R

(Figure 2) until the output distortion is

below some desired level. The output voltage swing can

then be set at any level by selecting R

Although the noise contribution of the linearizing diodes is

negligible relative to the contribution of the amplifier’s inter-

nal transistors, I

should be as large as possible. This mini-

mizes the dynamic junction resistance of the diodes (r

) and

maximizes their linearizing action when balanced against

. A value of 1 mA is recommended for I

unless the

specific application demands otherwise.

00798108

FIGURE 1. Linearizing Diodes

00798109

FIGURE 2. Voltage Controlled Amplifier

LM13700

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Applications

Voltage Controlled Amplifiers (Continued)

Stereo Volume Control

The circuit of Figure 4 uses the excellent matching of the two

LM13700 amplifiers to provide a Stereo Volume Control with

a typical channel-to-channel gain tracking of 0.3 dB. R

provided to minimize the output offset voltage and may be

replaced with two 510Ωresistors in AC-coupled applications.

For the component values given, amplifier gain is derived for

Figure 2 as being:

If V

is derived from a second signal source then the circuit

becomes an amplitude modulator or two-quadrant multiplier

as shown in Figure 5, where:

The constant term in the above equation may be cancelled

by feeding I

/2(V− + 1.4V) into I

. The circuit of

Figure 6 adds R

to provide this current, resulting in a

four-quadrant multiplier where R

is trimmed such that V

0V for V

IN2

=0V.R

also serves as the load resistor for I

00798110

FIGURE 3. Equivalent VCA Input Circuit

LM13700

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Stereo Volume Control (Continued)

00798111

FIGURE 4. Stereo Volume Control

00798112

FIGURE 5. Amplitude Modulator

LM13700

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Stereo Volume Control (Continued)

Noting that the gain of the LM13700 amplifier of Figure 3

may be controlled by varying the linearizing diode current I

as well as by varying I

ABC

,Figure 7 shows an AGC Amplifier

using this approach. As V

reaches a high enough amplitude

(3V

) to turn on the Darlington transistors and the lineariz-

ing diodes, the increase in I

reduces the amplifier gain so

as to hold V

at that level.

Voltage Controlled Resistors

An Operational Transconductance Amplifier (OTA) may be

used to implement a Voltage Controlled Resistor as shown in

Figure 8. A signal voltage applied at R

generates a V

the LM13700 which is then multiplied by the g

of the

amplifier to produce an output current, thus:

where g

≈19.2I

ABC

at 25˚C. Note that the attenuation of V

by R and R

is necessary to maintain V

within the linear

range of the LM13700 input.

Figure 9 shows a similar VCR where the linearizing diodes

are added, essentially improving the noise performance of

the resistor. A floating VCR is shown in Figure 10, where

each “end” of the “resistor” may be at any voltage within the

output voltage range of the LM13700.

00798113

FIGURE 6. Four-Quadrant Multiplier

00798114

FIGURE 7. AGC Amplifier

LM13700

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Voltage Controlled Resistors (Continued)

Voltage Controlled Filters

OTA’s are extremely useful for implementing voltage con-

trolled filters, with the LM13700 having the advantage that

the required buffers are included on the I.C. The VC Lo-Pass

Filter of Figure 11 performs as a unity-gain buffer amplifier at

frequencies below cut-off, with the cut-off frequency being

the point at which X

equals the closed-loop gain of

(R/R

). At frequencies above cut-off the circuit provides a

single RC roll-off (6 dB per octave) of the input signal ampli-

tude with a −3 dB point defined by the given equation, where

is again 19.2 x I

ABC

at room temperature. Figure 12

shows a VC High-Pass Filter which operates in much the

same manner, providing a single RC roll-off below the de-

fined cut-off frequency.

Additional amplifiers may be used to implement higher order

filters as demonstrated by the two-pole Butterworth Lo-Pass

Filter of Figure 13 and the state variable filter of Figure 14.

Due to the excellent g

tracking of the two amplifiers, these

filters perform well over several decades of frequency.

00798115

FIGURE 8. Voltage Controlled Resistor, Single-Ended

00798116

FIGURE 9. Voltage Controlled Resistor with Linearizing Diodes

LM13700

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Voltage Controlled Filters (Continued)

00798117

FIGURE 10. Floating Voltage Controlled Resistor

00798118

FIGURE 11. Voltage Controlled Low-Pass Filter

LM13700

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Voltage Controlled Filters (Continued)

00798119

FIGURE 12. Voltage Controlled Hi-Pass Filter

00798120

FIGURE 13. Voltage Controlled 2-Pole Butterworth Lo-Pass Filter

LM13700

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Voltage Controlled Filters (Continued)

Voltage Controlled Oscillators

The classic Triangular/Square Wave VCO of Figure 15 is

one of a variety of Voltage Controlled Oscillators which may

be built utilizing the LM13700. With the component values

shown, this oscillator provides signals from 200 kHz to below

2HzasI

is varied from 1 mA to 10 nA. The output

amplitudes are set by I

. Note that the peak differential

input voltage must be less than 5V to prevent zenering the

inputs.

A few modifications to this circuit produce the ramp/pulse

VCO of Figure 16. When V

is high, I

is added to I

increase amplifier A1’s bias current and thus to increase the

charging rate of capacitor C. When V

is low, I

goes to

zero and the capacitor discharge current is set by I

The VC Lo-Pass Filter of Figure 11 may be used to produce

a high-quality sinusoidal VCO. The circuit of Figure 16 em-

ploys two LM13700 packages, with three of the amplifiers

configured as lo-pass filters and the fourth as a limiter/

inverter. The circuit oscillates at the frequency at which the

loop phase-shift is 360˚ or 180˚ for the inverter and 60˚ per

filter stage. This VCO operates from 5 Hz to 50 kHz with less

than 1% THD.

00798121

FIGURE 14. Voltage Controlled State Variable Filter

LM13700

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Voltage Controlled Oscillators (Continued)

00798122

FIGURE 15. Triangular/Square-Wave VCO

00798123

FIGURE 16. Ramp/Pulse VCO

LM13700

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Voltage Controlled Oscillators (Continued)

Additional Applications

Figure 19 presents an interesting one-shot which draws no

power supply current until it is triggered. A positive-going

trigger pulse of at least 2V amplitude turns on the amplifier

through R

and pulls the non-inverting input high. The am-

plifier regenerates and latches its output high until capacitor

C charges to the voltage level on the non-inverting input. The

output then switches low, turning off the amplifier and dis-

charging the capacitor. The capacitor discharge rate is

speeded up by shorting the diode bias pin to the inverting

input so that an additional discharge current flows through D

when the amplifier output switches low. A special feature of

this timer is that the other amplifier, when biased from V

can perform another function and draw zero stand-by power

as well.

00798124

FIGURE 17. Sinusoidal VCO

00798125

Figure 18 shows how to build a VCO using one amplifier when the other

amplifier is needed for another function.

FIGURE 18. Single Amplifier VCO

LM13700

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Additional Applications (Continued)

The operation of the multiplexer of Figure 20 is very straight-

forward. When A1 is turned on it holds V

equal to V

IN1

and

when A2 is supplied with bias current then it controls V

and R

serve to stabilize the unity-gain configuration of

amplifiers A1 and A2. The maximum clock rate is limited to

about 200 kHz by the LM13700 slew rate into 150 pF when

the (V

IN1

–V

IN2

) differential is at its maximum allowable value

of 5V.

The Phase-Locked Loop of Figure 21 uses the four-quadrant

multiplier of Figure 6 and the VCO of Figure 18 to produce a

PLL with a ±5% hold-in range and an input sensitivity of

about 300 mV.

00798126

FIGURE 19. Zero Stand-By Power Timer

00798127

FIGURE 20. Multiplexer

LM13700

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Additional Applications (Continued)

The Schmitt Trigger of Figure 22 uses the amplifier output

current into R to set the hysteresis of the comparator; thus

=2xRxI

. Varying I

will produce a Schmitt Trigger with

variable hysteresis.

Figure 23 shows a Tachometer or Frequency-to-Voltage con-

verter. Whenever A1 is toggled by a positive-going input, an

amount of charge equal to (V

–V

is sourced into C

and

. This once per cycle charge is then balanced by the

current of V

. The maximum F

is limited by the amount

of time required to charge C

from V

to V

with a current of

, where V

and V

represent the maximum low and maxi-

mum high output voltage swing of the LM13700. D1 is added

to provide a discharge path for C

when A1 switches low.

The Peak Detector of Figure 24 uses A2 to turn on A1

whenever V

becomes more positive than V

. A1 then

charges storage capacitor C to hold V

equal to V

PK.

Pulling the output of A2 low through D1 serves to turn off A1

so that V

remains constant.

00798128

FIGURE 21. Phase Lock Loop

00798129

FIGURE 22. Schmitt Trigger

LM13700

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Additional Applications (Continued)

The Ramp-and-Hold of Figure 26 sources I

into capacitor C

whenever the input to A1 is brought high, giving a ramp-rate

of about 1V/ms for the component values shown.

The true-RMS converter of Figure 27 is essentially an auto-

matic gain control amplifier which adjusts its gain such that

the AC power at the output of amplifier A1 is constant. The

output power of amplifier A1 is monitored by squaring ampli-

fier A2 and the average compared to a reference voltage

with amplifier A3. The output of A3 provides bias current to

the diodes of A1 to attenuate the input signal. Because the

output power of A1 is held constant, the RMS value is

constant and the attenuation is directly proportional to the

RMS value of the input voltage. The attenuation is also

proportional to the diode bias current. Amplifier A4 adjusts

the ratio of currents through the diodes to be equal and

therefore the voltage at the output of A4 is proportional to the

RMS value of the input voltage. The calibration potentiom-

eter is set such that V

reads directly in RMS volts.

00798130

FIGURE 23. Tachometer

00798131

FIGURE 24. Peak Detector and Hold Circuit

LM13700

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Additional Applications (Continued)

00798132

FIGURE 25. Sample-Hold Circuit

00798133

FIGURE 26. Ramp and Hold

LM13700

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Additional Applications (Continued)

The circuit of Figure 28 is a voltage reference of variable

Temperature Coefficient. The 100 kΩpotentiometer adjusts

the output voltage which has a positive TC above 1.2V, zero

TC at about 1.2V, and negative TC below 1.2V. This is

accomplished by balancing the TC of the A2 transfer function

against the complementary TC of D1.

The wide dynamic range of the LM13700 allows easy control

of the output pulse width in the Pulse Width Modulator of

Figure 29.

For generating I

ABC

over a range of 4 to 6 decades of

current, the system of Figure 30 provides a logarithmic cur-

rent out for a linear voltage in.

Since the closed-loop configuration ensures that the input to

A2 is held equal to 0V, the output current of A1 is equal to

=−V

The differential voltage between Q1 and Q2 is attenuated by

the R1,R2 network so that A1 may be assumed to be oper-

ating within its linear range. From Equation (5), the input

voltage to A1 is:

The voltage on the base of Q1 is then

The ratio of the Q1 and Q2 collector currents is defined by:

Combining and solving for I

ABC

yields:

This logarithmic current can be used to bias the circuit of

Figure 4 to provide temperature independent stereo attenu-

ation characteristic.

00798134

FIGURE 27. True RMS Converter

LM13700

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Additional Applications (Continued)

00798135

FIGURE 28. Delta VBE Reference

00798136

FIGURE 29. Pulse Width Modulator

LM13700

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Additional Applications (Continued)

00798137

FIGURE 30. Logarithmic Current Source

LM13700

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Physical Dimensions inches (millimeters) unless otherwise noted

S.O. Package (M)

Order Number LM13700M or LM13700MX

NS Package Number M16A

Molded Dual-In-Line Package (N)

Order Number LM13700N

NS Package Number N16A

LM13700

www.national.com25

Notes

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COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

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www.national.com

LM13700 Dual Operational Transconductance Amplifiers with Linearizing Diodes and Buffers

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.

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