Sample & Buy Product Folder Support & Community Tools & Software Technical Documents LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 LM3429/-Q1 N-Channel Controller for Constant Current LED Drivers 1 Features 3 Description * The LM3429 is a versatile high voltage N-channel MosFET controller for LED drivers. It can be easily configured in buck, boost, buck-boost and SEPIC topologies. This flexibility, along with an input voltage rating of 75V, makes the LM3429 ideal for illuminating LEDs in a very diverse, large family of applications. 1 * * * * * * * * * * * * LM3429-Q1 is AEC-Q100 Grade 1 Qualified for Automotive Applications VIN Range From 4.5 V to 75 V Adjustable Current Sense Voltage High-Side Current Sensing 2-, 1-A Peak MosFET Gate Driver Input Undervoltage Protection Overvoltage Protection PWM Dimming Analog Dimming Cycle-by-Cycle Current Limit Programmable Switching Frequency Low Profile 14-lead HTSSOP Package Thermal Shutdown Adjustable high-side current sense voltage allows for tight regulation of the LED current with the highest efficiency possible. The LM3429 uses Predictive Offtime (PRO) control, which is a combination of peak current-mode control and a predictive off-timer. This method of control eases the design of loop compensation while providing inherent input voltage feed-forward compensation. The LM3429 includes a high-voltage startup regulator that operates over a wide input range of 4.5 V to 75 V. The internal PWM controller is designed for adjustable switching frequencies of up to 2 MHz, thus enabling compact solutions. Additional features include analog dimming, PWM dimming, overvoltage protection, undervoltage lock-out, cycle-by-cycle current limit, and thermal shutdown. 2 Applications * * * * * LED Drivers - Buck, Boost, Buck-Boost, SEPIC Indoor and Outdoor SSL Automotive General Illumination Constant-Current Regulators Device Information(1) PART NUMBER LM3429 LM3429-Q1 PACKAGE BODY SIZE (NOM) HTSSOP (14) 5.00 mm x 4.40 mm (1) For all available packages, see the orderable addendum at the end of the data sheet. Typical Boost Application Circuit VIN 1 2 3 4 5 6 VIN LM3429 HSN COMP HSP CSH IS RCT VCC AGND GATE OVP PGND 14 13 12 11 ILED 10 9 DAP PWM 7 nDIM NC 8 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Table of Contents 1 2 3 4 5 6 7 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 1 1 1 2 3 4 6.1 6.2 6.3 6.4 6.5 6.6 4 4 4 5 5 7 Absolute Maximum Ratings ...................................... ESD Ratings.............................................................. Recommended Operating Conditions....................... Thermal Information .................................................. Electrical Characteristics .......................................... Typical Characteristics ............................................. Detailed Description .............................................. 9 7.1 7.2 7.3 7.4 Overview ................................................................... 9 Functional Block Diagram ......................................... 9 Feature Description................................................. 10 Device Functional Modes........................................ 21 8 Application and Implementation ........................ 22 8.1 Application Information............................................ 22 8.2 Typical Applications ................................................ 24 9 Power Supply Recommendations...................... 53 9.1 Input Supply Current Limit ...................................... 53 10 Layout................................................................... 53 10.1 Layout Guidelines ................................................. 53 10.2 Layout Example .................................................... 54 11 Device and Documentation Support ................. 55 11.1 11.2 11.3 11.4 11.5 11.6 11.7 Device Support...................................................... Documentation Support ........................................ Related Links ........................................................ Community Resources.......................................... Trademarks ........................................................... Electrostatic Discharge Caution ............................ Glossary ................................................................ 55 55 55 55 55 56 56 12 Mechanical, Packaging, and Orderable Information ........................................................... 56 4 Revision History NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision G (April 2013) to Revision H * Page Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information section ................................................................................................................................................................................... 1 Changes from Revision F (May 2013) to Revision G Page * Changed layout of National Data Sheet to TI format ........................................................................................................... 51 * Changed layout of National Data Sheet to TI format ........................................................................................................... 52 2 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 5 Pin Configuration and Functions PWP Package 14- Pin HTSSOP Top View VIN 1 14 HSN COMP 2 13 HSP CSH 3 RCT 12 IS DAP 15 4 AGND 5 11 VCC 10 GATE OVP 6 9 PGND nDIM 7 8 NC Pin Functions PIN NO. NAME I/O DESCRIPTION APPLICATION INFORMATION 1 VIN I Input Voltage Bypass with 100 nF capacitor to AGND as close to the device as possible in the circuit board layout. 2 COMP I Compensation Connect a capacitor to AGND to set compensation. 3 CSH I Current Sense High 4 RCT I Resistor Capacitor Timing 5 AGND GND Analog Ground 6 OVP I Overvoltage Protection 7 nDIM I Not DIM input Connect a PWM signal for dimming as detailed in the PWM Dimming section and/or a resistor divider from VIN to program input undervoltage lockout (UVLO). Turn-on threshold is 1.24 V and hysteresis for turn-off is provided by 20 A current source. 8 NC No Connection Leave open. 9 PGND GND Power Ground Connect to AGND through DAP copper pad to provide ground return for GATE. 10 GATE O Gate Drive Output 11 VCC I Internal Regulator Output 12 IS I Main Switch Current Sense Connect to the drain of the main N-channel MosFET switch for RDSON sensing or to a sense resistor installed in the source of the same device. 13 HSP I LED Current Sense Positive Connect through a series resistor to LED current sense resistor (positive). 14 HSN I LED Current Sense Negative Connect through a series resistor to LED current sense resistor (negative). DAP (15) DAP GND Thermal pad on bottom of IC Connect to AGND and PGND. Connect a resistor to AGND to set signal current. For analog dimming, connect current source or potentiometer to AGND (see Analog Dimming section). Connect a resistor from the switch node and a capacitor to AGND to set the switching frequency. Connect to PGND through the DAP copper circuit board pad to provide proper ground return for CSH, COMP, and RCT. Connect to a resistor divider from the output (VO) or the input to program output overvoltage lockout (OVLO). Turn-off threshold is 1.24 V and hysteresis for turn-on is provided by 20 A current source. Connect to the gate of the external NFET. Bypass with a 2.2 F-3.3 F, ceramic capacitor to PGND. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 3 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings over operating free-air temperature range (unless otherwise noted) (1) (2) MIN MAX VIN, nDIM -0.3 76 OVP, HSP, HSN -0.3 76 RCT -0.3 3 -0.3 76 IS Voltage -2 for 100 ns VCC -0.3 8 COMP, CSH -0.3 6 GATE -0.3 VCC VCC+2.5 for 100 ns -0.3 0.3 -2.5 2.5 for 100 ns VIN, nDIM OVP, HSP, HSN RCT -1 A mA -200 200 A -1 1 mA 260 C 150 C Maximum Junction Temperature Internally Limited Maximum Lead Temperature (Reflow and Solder) (3) Continuous Power Dissipation Internally Limited Storage Temperature, Tstg (3) mA -1 GATE (2) -1 -100 5 IS COMP, CSH (1) V -2.5 for 100 ns PGND Continuous Current UNIT -65 Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/Distributors for availability and specifications. Refer to http://www.ti.com/packaging for more detailed information and mounting techniques. 6.2 ESD Ratings VALUE UNIT LM3429 IN PWP PACKAGE V(ESD) Electrostatic discharge Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins (1) 2000 Charged device model (CDM), per JEDEC specification JESD22-C101, all pins (2) 1000 Human body model (HBM), per AEC Q100-002 (3) 2000 Charged device model (CDM), per AEC Q100-011 1000 V LM3429-Q1 IN PWP PACKAGE V(ESD) (1) (2) (3) Electrostatic discharge V JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification. 6.3 Recommended Operating Conditions MIN MAX UNIT Operating Junction Temperature Range -40 125 C Input Voltage VIN 4.5 75 V 4 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 6.4 Thermal Information THERMAL METRIC (1) LM3429-Q1 LM3429 PWP (HTSSOP) PWP (HTSSOP) 14 PINS 14 PINS UNIT RJA Junction-to-ambient thermal resistance 47.8 47.8 C/W RJC(top) Junction-to-case (top) thermal resistance 26.5 26.5 C/W RJB Junction-to-board thermal resistance 22.3 22.3 C/W JT Junction-to-top characterization parameter 0.7 0.7 C/W JB Junction-to-board characterization parameter 22.1 22.1 C/W RJC(bot) Junction-to-case (bottom) thermal resistance 3.3 3.3 C/W (1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application report, SPRA953. 6.5 Electrical Characteristics MIN and MAX limits apply TJ = (-40C to 125C) unless specified otherwise. Unless otherwise stated the following condition applies: VIN = 14 V. PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) 6.9 7.35 UNIT STARTUP REGULATOR (VCC) VCC-REG VCC Regulation ICC = 0 mA 6.3 ICC-LIM VCC Current Limit VCC = 0V 20 IQ Quiescent Current Static VCC-UVLO VCC UVLO Threshold VCC Increasing VCC-HYS VCC UVLO Hysteresis VCC Decreasing 3.7 V 27 1.6 3 4.17 4.5 mA 4.08 V 0.1 OVERVOLTAGE PROTECTION (OVP) VTH-OVP OVP OVLO Threshold OVP Increasing IHYS-OVP OVP Hysteresis Source Current OVP Active (high) 1.18 1.24 1.28 V 10 20 30 A 1.235 1.26 V ERROR AMPLIFIER VCSH CSH Reference Voltage With Respect to AGND 1.21 Error Amplifier Input Bias Current MIN, MAX: TJ = 25C -0.6 0 0.6 10 26 40 COMP Sink / Source Current Transconductance (3) Linear Input Range Transconductance Bandwidth -6dB Unloaded Response (3), MIN: TJ = 25C 0.5 A 100 A/V 125 mV 1 MHz OFF TIMER (RCT) tOFF-MIN Minimum Off-time RRCT RCT Reset Pulldown Resistance RCT = 1V through 1 k VRCT VIN/25 Reference Voltage VIN = 14V 35 75 ns 36 120 540 565 585 mV 700 800 900 mV 215 245 275 mV 35 75 250 450 PWM COMPARATOR COMP to PWM Offset CURRENT LIMIT (IS) VLIM Current Limit Threshold VLIM Delay to Output tON-MIN (1) (2) (3) Leading Edge Blanking Time 75 ns All limits specified at room temperature (TYP) and at temperature extremes (MIN/MAX). All room temperature limits are 100% production tested. All limits at temperature extremes are specified through correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL). Typical numbers are at 25C and represent the most likely norm. These electrical parameters are specified by design, and are not verified by test. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 5 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Electrical Characteristics (continued) MIN and MAX limits apply TJ = (-40C to 125C) unless specified otherwise. Unless otherwise stated the following condition applies: VIN = 14 V. PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) 10 A 20 119 mA/V -1.5 0 1.5 A -7 0 7 mV 250 500 UNIT HIGH SIDE TRANSCONDUCTANCE AMPLIFIER Input Bias Current Transconductance Input Offset Current Input Offset Voltage Transconductance Bandwidth ICSH = 100 A (3), MIN: TJ = 25C kHz GATE DRIVER (GATE) RSRC(GATE) GATE Sourcing Resistance GATE = High 2 6 RSNK(GATE) GATE Sinking Resistance GATE = Low 1.3 4.5 UNDERVOLTAGE LOCKOUT and DIM INPUT (nDIM) VTH-nDIM nDIM / UVLO Threshold 1.18 1.24 1.28 V IHYS-nDIM nDIM Hysteresis Current 10 20 30 A THERMAL SHUTDOWN TSD Thermal Shutdown Threshold (3) 165 THYS Thermal Shutdown Hysteresis (3) 25 6 Submit Documentation Feedback C Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 6.6 Typical Characteristics TA= 25C and VIN = 14 V unless otherwise specified. The measurements for Figure 1 and Figure 3 were made using the standard boost evaluation board from AN-1986 (SNVA404). The measurements for Figure 2, Figure 4, and Figure 5, Figure 6 were made using the standard buck-boost evaluation board from AN-1985 (SNVA403). 100 100 95 EFFICIENCY (%) EFFICIENCY (%) 95 90 90 85 80 85 75 70 80 VIN (V) 32 48 VIN (V) Figure 1. Boost Efficiency vs Input Voltage VO = 32 V (9 LEDs) Figure 2. Buck-Boost Efficiency vs Input Voltage VO = 20 V (6 LEDs) 15 20 25 0 30 1.00 1.05 0.99 1.03 ILED (A) ILED (A) 10 0.98 16 64 80 1.01 0.99 0.97 0.97 0.96 5 10 15 20 25 0 30 16 32 48 64 80 VIN (V) Figure 3. Boost LED Current vs Input Voltage VO = 32 V (9 LEDs) Figure 4. Buck-boost LED Current vs Input Voltage VO = 20 V (6 LEDs) 1.0 1.0 0.8 0.8 0.6 0.6 ILED (A) ILED (A) VIN (V) 0.4 0.2 500 Hz 0.4 100 Hz 0.2 0.0 0 20 40 60 80 100 0.0 0 ICSH (eA) 20 40 60 80 100 DUTY CYCLE (%) Figure 5. Analog Dimming VO = 20 V (6 LEDs) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Figure 6. PWM Dimming VO = 20V (6 LEDs) Submit Documentation Feedback 7 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Characteristics (continued) 1.250 7.10 1.245 7.05 1.240 7.00 VCC (V) VCSH (V) TA= 25C and VIN = 14 V unless otherwise specified. The measurements for Figure 1 and Figure 3 were made using the standard boost evaluation board from AN-1986 (SNVA404). The measurements for Figure 2, Figure 4, and Figure 5, Figure 6 were made using the standard buck-boost evaluation board from AN-1985 (SNVA403). 1.235 6.95 1.230 6.90 1.225 6.85 1.220 6.80 -50 -14 22 58 94 130 -50 -14 TEMPERATURE (C) Figure 7. VCSH vs. Junction Temperature 58 94 130 Figure 8. VCC vs. Junction Temperature 246 569 568 244 VLIM (mV) VRCT (mV) 22 TEMPERATURE (C) 567 566 242 240 565 564 -50 238 -14 22 58 94 -50 130 -14 22 58 94 130 TEMPERATURE (C) TEMPERATURE (C) Figure 10. VLIM vs. Junction Temperature Figure 9. VRCT vs. Junction Temperature 280 tON-MIN (ns) 275 270 265 260 255 250 -50 -14 22 58 94 130 TEMPERATURE (C) Figure 11. tON-MIN vs. Junction Temperature 8 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 7 Detailed Description 7.1 Overview The LM3429 is an N-channel MosFET (NFET) controller for buck, boost and buck-boost current regulators which are ideal for driving LED loads. The controller has wide input voltage range allowing for regulation of a variety of LED loads. The high-side differential current sense, with low adjustable threshold voltage, provides an excellent method for regulating output current while maintaining high system efficiency. The LM3429 uses a Predictive Offtime (PRO) control architecture that allows the regulator to be operated using minimal external control loop compensation, while providing an inherent cycle-by-cycle current limit. The adjustable current sense threshold provides the capability to amplitude (analog) dim the LED current and the output enable/disable function allows for PWM dimming using no external components. When designing, the maximum attainable LED current is not internally limited because the LM3429 is a controller. Instead it is a function of the system operating point, component choices, and switching frequency allowing the LM3429 to easily provide constant currents up to 5A. This simple controller contains all the features necessary to implement a high-efficiency versatile LED driver. 7.2 Functional Block Diagram VIN 6.9V LDO Regulator UVLO UVLO HYSTERESIS VccUVLO Standby REFERENCE 1.24V TLIM Thermal Limit Dimming 20 PA nDIM VCC 1.24V VIN/25 Start new on time Reset Dominant Q GATE R QB PGND S RCT VCC COMP OVP HYSTERESIS PWM 1.24V CSH 20 PA OVP 1.24V 800 mV HSP LOGIC HSN CURRENT LIMIT AGND IS 245 mV LEB Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 9 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 7.3 Feature Description 7.3.1 Current Regulators Current regulators can be designed to accomplish three basic functions: buck, boost, and buck-boost. All three topologies in their most basic form contain a main switching MosFET, a recirculating diode, an inductor and capacitors. The LM3429 is designed to drive a ground referenced NFET which is perfect for a standard boost regulator. Buck and buck-boost regulators, on the other hand, usually have a high-side switch. When driving an LED load, a ground referenced load is often not necessary, therefore a ground referenced switch can be used to drive a floating load instead. The LM3429 can then be used to drive all three basic topologies as shown in the Typical Applications section. Looking at the buck-boost design, the basic operation of a current regulator can be analyzed. During the time that the NFET (Q1) is turned on (tON), the input voltage source stores energy in the inductor (L1) while the output capacitor (CO) provides energy to the LED load. When Q1 is turned off (tOFF), the re-circulating diode (D1) becomes forward biased and L1 provides energy to both CO and the LED load. Figure 12 shows the inductor current (iL(t)) waveform for a regulator operating in CCM. iL (t) IL-MAX AiL-PP IL IL-MIN tON = DTS tOFF = (1-D)TS t 0 TS Figure 12. Ideal CCM Regulator Inductor Current iL(t) The average output LED current (ILED) is proportional to the average inductor current (IL) , therefore if IL is tightly controlled, ILED will be well regulated. As the system changes input voltage or output voltage, the ideal duty cycle (D) is varied to regulate IL and ultimately ILED. For any current regulator, D is a function of the conversion ratio: Buck D= VO VIN (1) VO - VIN VO (2) Boost D= Buck-Boost D= VO VO + VIN (3) 7.3.2 Predictive Off-Time (PRO) Control PRO control is used by the LM3429 to control ILED. It is a combination of average peak current control and a oneshot off-timer that varies with input voltage. The LM3429 uses peak current control to regulate the average LED current through an array of HBLEDs. This method of control uses a series resistor in the LED path to sense LED current and can use either a series resistor in the MosFET path or the MosFET RDS-ON for both cycle-by-cycle current limit and input voltage feed forward. D is indirectly controlled by changes in both tOFF and tON, which vary depending on the operating point. 10 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Feature Description (continued) Even though the off-time control is quasi-hysteretic, the input voltage proportionality in the off-timer creates an essentially constant switching frequency over the entire operating range for boost and buck-boost topologies. The buck topology can be designed to give constant ripple over either input voltage or output voltage, however switching frequency is only constant at a specific operating point . This type of control minimizes the control loop compensation necessary in many switching regulators, simplifying the design process. The averaging mechanism in the peak detection control loop provides extremely accurate LED current regulation over the entire operating range. PRO control was designed to mitigate "current mode instability" (also called "sub-harmonic oscillation") found in standard peak current mode control when operating near or above 50% duty cycles. When using standard peak current mode control with a fixed switching frequency, this condition is present, regardless of the topology. However, using a constant off-time approach, current mode instability cannot occur, enabling easier design and control. Predictive off-time advantages: * There is no current mode instability at any duty cycle. * Higher duty cycles / voltage transformation ratios are possible, especially in the boost regulator. The only disadvantage is that synchronization to an external reference frequency is generally not available. 7.3.3 Switching Frequency An external resistor (RT) connected between the RCT pin and the switch node (where D1, Q1, and L1 connect), in combination with a capacitor (CT) between the RCT and AGND pins, sets the off-time (tOFF) as shown in Figure 13. For boost and buck-boost topologies, the VIN proportionality ensures a virtually constant switching frequency (fSW). VSW LM3429 RT VIN/25 Start tON RCT CT Reset timer Figure 13. Off-timer Circuitry for Boost and Buck-boost Regulators For a buck topology, RT and CT are also used to set tOFF, however the VIN proportionality will not ensure a constant switching frequency. Instead, constant ripple operation can be achieved. Changing the connection of RT in Figure 13 from VSW to VIN will provide a constant ripple over varying VIN. Adding a PNP transistor as shown in Figure 14 will provide constant ripple over varying VO. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 11 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Feature Description (continued) VIN RSNS RT LM3429 VIN/25 LED- Start tON RCT CT Reset timer Figure 14. Off-timer Circuitry for Buck Regulators The switching frequency is defined: Buck (Constant Ripple vs. VIN) fSW = 25 x ( VIN - VO ) RT x CT X VIN (4) Buck (Constant Ripple vs. VO) 25 x (VIN x VO - VO ) 2 fSW = 2 RT x C T x VIN (5) Boost and Buck-Boost fSW = 25 R T x CT (6) For all topologies, the CT capacitor is recommended to be 1 nF and should be located very close to the LM3429. 7.3.4 Average LED Current The LM3429 uses an external current sense resistor (RSNS) placed in series with the LED load to convert the LED current (ILED) into a voltage (VSNS) as shown in Figure 15. The HSP and HSN pins are the inputs to the high-side sense amplifier which are forced to be equal potential (VHSP=VHSN) through negative feedback. Because of this, the VSNS voltage is forced across RHSP to generate the signal current (ICSH) which flows out of the CSH pin and through the RCSH resistor. The error amplifier will regulate the CSH pin to 1.24 V, therefore ICSH can be calculated: ICSH = VSNS RHSP (7) This means VSNS will be regulated as follows: RHSP VSNS = 1.24V x RCSH (8) ILED can then be calculated: 12 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Feature Description (continued) ILED = VSNS 1.24V RHSP x = RSNS RSNS RCSH (9) The selection of the three resistors (RSNS, RCSH, and RHSP) is not arbitrary. For matching and noise performance, the suggested signal current ICSH is approximately 100 A. This current does not flow in the LEDs and will not affect either the off state LED current or the regulated LED current. ICSH can be above or below this value, but the high-side amplifier offset characteristics may be affected slightly. In addition, to minimize the effect of the high-side amplifier voltage offset on LED current accuracy, the minimum VSNS is suggested to be 50 mV. Finally, a resistor (RHSN = RHSP) should be placed in series with the HSN pin to cancel out the effects of the input bias current (~10 A) of both inputs of the high-side sense amplifier. The CSH pin can also be used as a low-side current sense input regulated to 1.24 V. The high-side sense amplifier is disabled if HSP and HSN are tied to GND. LM3429 ILED RHSP HSP High-Side Sense Amplifier ICSH VSNS RSNS RHSN HSN RCSH CSH Error Amplifier 1.24V CCMP To PWM Comparator COMP Figure 15. LED Current Sense Circuitry 7.3.5 Analog Dimming The CSH pin can be used to analog dim the LED current by adjusting the current sense voltage (VSNS). There are several different methods to adjust VSNS using the CSH pin: 1. External variable resistance : Adjust a potentiometer placed in series with RCSH to vary VSNS. 2. External variable current source: Source current (0 A to ICSH) into the CSH pin to adjust VSNS. In general, analog dimming applications require a lower switching frequency to minimize the effect of the leading edge blanking circuit. As the LED current is reduced, the output voltage and the duty cycle decreases. Eventually, the minimum on-time is reached. The lower the switching frequency, the wider the linear dimming range. Figure 16 shows how both methods are physically implemented. Method 1 uses an external potentiometer in the CSH path which is a simple addition to the existing circuitry. However, the LEDs cannot dim completely because there is always some resistance causing signal current to flow. This method is also susceptible to noise coupling at the CSH pin because the potentiometer increases the size of the signal current loop. Method 2 provides a complete dimming range and better noise performance, though it is more complex. It consists of a PNP current mirror and a bias network consisting of an NPN, 2 resistors and a potentiometer (RADJ), where RADJ controls the amount of current sourced into the CSH pin. A higher resistance value will source more current into the CSH pin causing less regulated signal current through RHSP, effectively dimming the LEDs. VREF should be a precise external voltage reference, while Q7 and Q8 should be a dual pair PNP for best matching and performance. The additional current (IADD) sourced into the CSH pin can be calculated: Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 13 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Feature Description (continued) IADD = RADJ x VREF * R + R - VBE-Q6 (c) ADJ MAX RBIAS (10) The corresponding ILED for a specific IADD is: RHSP* ILED = ICSH - IADD x RSNS ( ) (c) (11) Variable Current Source VCC LM3429 VREF Q8 Q7 RMAX Q6 RADJ RBIAS CSH RCSH RADJ Variable Resistance Figure 16. Analog Dimming Circuitry 7.3.6 Current Sense and Current Limit The LM3429 achieves peak current mode control using a comparator that monitors the MosFET transistor current, comparing it with the COMP pin voltage as shown in Figure 17. Further, it incorporates a cycle-by-cycle overcurrent protection function. Current limit is accomplished by a redundant internal current sense comparator. If the voltage at the current sense comparator input (IS) exceeds 245 mV (typical), the on cycle is immediately terminated. The IS input pin has an internal N-channel MosFET which pulls it down at the conclusion of every cycle. The discharge device remains on an additional 250 ns (typical) after the beginning of a new cycle to blank the leading edge spike on the current sense signal. The leading edge blanking (LEB) determines the minimum achievable on-time (tON-MIN). 14 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Feature Description (continued) LM3429 COMP Q1 GATE PWM 800 mV IS 245 mV IT RLIM LEB PGND Figure 17. Current Sense / Current Limit Circuitry There are two possible methods to sense the transistor current. The RDS-ON of the main power MosFET can be used as the current sense resistance because the IS pin was designed to withstand the high voltages present on the drain when the MosFET is in the off state. Alternatively, a sense resistor located in the source of the MosFET may be used for current sensing, however a low inductance (ESL) type is suggested. The cycle-by-cycle current limit (ILIM) can be calulated using either method as the limiting resistance (RLIM): ILIM = 245 mV RLIM (12) In general, the external series resistor allows for more design flexibility, however it is important to ensure all of the noise sensitive low power ground connections are connected together local to the controller and a single connection is made to the high current PGND (sense resistor ground point). 7.3.7 Control Loop Compensation The LM3429 control loop is modeled like any current mode controller. Using a first order approximation, the uncompensated loop can be modeled as a single pole created by the output capacitor and, in the boost and buck-boost topologies, a right half plane zero created by the inductor, where both have a dependence on the LED string dynamic resistance. There is also a high frequency pole in the model, however it is above the switching frequency and plays no part in the compensation design process therefore it will be neglected. Because ceramic capacitance is recommended for use with LED drivers due to long lifetimes and high ripple current rating, the ESR of the output capacitor can also be neglected in the loop analysis. Finally, there is a DC gain of the uncompensated loop which is dependent on internal controller gains and the external sensing network. A buck-boost regulator will be used as an example case. See the Typical Applications section for compensation of all topologies. The uncompensated loop gain for a buck-boost regulator is given by the following equation: s * 1 ZZ1 (c) TU = TU0 x s * 1+ ZP1 (c) (13) Where the uncompensated DC loop gain of the system is described as: Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 15 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Feature Description (continued) TU0 = Dc x 500V x RCSH x RSNS (1+ D) x RHSP x R LIM = Dc x 620V (1+ D) x ILED x R LIM (14) 3 And the output pole (P1) is approximated: 1+ D ZP1 = rD x CO (15) And the right half plane zero (Z1) is: rD x Dc2 ZZ1 = D x L1 (16) 100 oZ1 80 135 oP1 90 GAIN GAIN (dB) 0 40 PHASE -45 20 0 Phase Margin -90 0 -20 -135 -40 -180 -60 1e-1 PHASE () 45 60 1e1 1e3 1e5 -225 1e7 FREQUENCY (Hz) Figure 18. Uncompensated Loop Gain Frequency Response Figure 18 shows the uncompensated loop gain in a worst-case scenario when the RHP zero is below the output pole. This occurs at high duty cycles when the regulator is trying to boost the output voltage significantly. The RHP zero adds 20dB/decade of gain while loosing 45/decade of phase which places the crossover frequency (when the gain is zero dB) extremely high because the gain only starts falling again due to the high frequency pole (not modeled or shown in figure). The phase will be below -180 at the crossover frequency which means there is no phase margin (180 + phase at crossover frequency) causing system instability. Even if the output pole is below the RHP zero, the phase will still reach -180 before the crossover frequency in most cases yielding instability. 16 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Feature Description (continued) LM3429 ILED RHSP HSP High-Side Sense Amplifier CFS VSNS RSNS RHSN HSN RFS sets oP3 RCSH Error Amplifier CSH 1.24V sets oP2 CCMP RO To PWM Comparator COMP Figure 19. Compensation Circuitry To mitigate this problem, a compensator should be designed to give adequate phase margin (above 45) at the crossover frequency. A simple compensator using a single capacitor at the COMP pin (CCMP) will add a dominant pole to the system, which will ensure adequate phase margin if placed low enough. At high duty cycles (as shown in Figure 18), the RHP zero places extreme limits on the achievable bandwidth with this type of compensation. However, because an LED driver is essentially free of output transients (except catastrophic failures open or short), the dominant pole approach, even with reduced bandwidth, is usually the best approach. The dominant compensation pole (P2) is determined by CCMP and the output resistance (RO) of the error amplifier (typically 5 M): 1 ZP2 6 5x10 : x CCMP (17) It may also be necessary to add one final pole at least one decade above the crossover frequency to attenuate switching noise and, in some cases, provide better gain margin. This pole can be placed across RSNS to filter the ESL of the sense resistor at the same time. Figure 19 shows how the compensation is physically implemented in the system. The high frequency pole (P3) can be calculated: 1 ZP3 = RFS x CFS (18) The total system transfer function becomes: s * 1 ZZ1 (c) T = TU0 x s * s * s * 1+ ZP1 x 1+ ZP2 x 1+ ZP3 (c) (c) (c) (19) The resulting compensated loop gain frequency response shown in Figure 20 indicates that the system has adequate phase margin (above 45) if the dominant compensation pole is placed low enough, ensuring stability: Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 17 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Feature Description (continued) 90 80 oP2 45 60 0 GAIN 20 0 oZ1 -90 PHASE oP3 -20 -40 -45 oP1 -135 60 Phase Margin -180 -225 -60 -80 1e-1 PHASE () GAIN (dB) 40 1e1 1e3 1e5 -270 1e7 FREQUENCY (Hz) Figure 20. Compensated Loop Gain Frequency Response 7.3.8 Output Overvoltage Lockout (OVLO) The LM3429 can be configured to detect an output (or input) overvoltage condition through the OVP pin. The pin features a precision 1.24-V threshold with 20 A (typical) of hysteresis current as shown in Figure 21. When the OVLO threshold is exceeded, the GATE pin is immediately pulled low and a 20 A current source provides hysteresis to the lower threshold of the OVLO hysteretic band. LM3429 VO 20 PA ROV2 OVP 1.24V OVLO ROV1 Figure 21. Overvoltage Protection Circuitry If the LEDs are referenced to a potential other than ground (floating), as in the buck-boost and buck configuration, the output voltage (VO) should be sensed and translated to ground by using a single PNP as shown in Figure 22. 18 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Feature Description (continued) LED+ ROV2 LM3429 LEDOVP ROV1 Figure 22. Floating Output OVP Circuitry The overvoltage turnoff threshold (VTURN-OFF) is defined as follows: Ground Referenced R + ROV 2* VTURN - OFF = 1.24V x OV1 (c) R OV1 (20) Floating 0.5 x R OV1+ R OV2* VTURN - OFF = 1.24V x R OV1 (c) (21) In the ground referenced configuration, the voltage across ROV2 is VO - 1.24 V whereas in the floating configuration it is VO - 620 mV where 620 mV approximates the VBE of the PNP transistor. The overvoltage hysteresis (VHYSO) is defined as follows: VHYSO = 20 PA x ROV2 (22) 7.3.9 Input Undervoltage Lockout (UVLO) The nDIM pin is a dual-function input that features an accurate 1.24 V threshold with programmable hysteresis as shown in Figure 23. This pin functions as both the PWM dimming input for the LEDs and as a VIN UVLO. When the pin voltage rises and exceeds the 1.24 V threshold, 20 A (typical) of current is driven out of the nDIM pin into the resistor divider providing programmable hysteresis. LM3429 VIN 20 PA RUV2 RUV1 nDIM 1.24V RUVH UVLO (optional) Figure 23. UVLO Circuit Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 19 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Feature Description (continued) When using the nDIM pin for UVLO and PWM dimming concurrently, the UVLO circuit can have an extra series resistor to set the hysteresis. This allows the standard resistor divider to have smaller resistor values minimizing PWM delays due to a pulldown MosFET at the nDIM pin (see PWM Dimming section). In general, at least 3V of hysteresis is necessary when PWM dimming if operating near the UVLO threshold. The turn-on threshold (VTURN-ON) is defined as follows: R UV1 + RUV2* VTURN ON - = 1. 24V x (c) RUV1 (23) The hysteresis (VHYS) is defined as follows: UVLO Only VHYS = 20 PA x RUV2 (24) PWM Dimming and UVLO R x (RUV1 + RUV2) * VHYS = 20 PA x RUV2 + UVH RUV1 (c) (25) 7.3.10 PWM Dimming The active low nDIM pin can be driven with a PWM signal which controls the main NFET (Q1). The brightness of the LEDs can be varied by modulating the duty cycle of this signal. LED brightness is approximately proportional to the PWM signal duty cycle, so 30% duty cycle equals approximately 30% LED brightness. This function can be ignored if PWM dimming is not required by using nDIM solely as a VIN UVLO input as described in the Input Undervoltage Lockout (UVLO) section or by tying it directly to VCC or VIN (if less than 76VDC). Inverted PWM VIN LM3429 DDIM RUV2 nDIM RUVH RUV1 QDIM Standard PWM Figure 24. PWM Dimming Circuit Figure 24 shows two ways the PWM signal can be applied to the nDIM pin: 1. Connect the dimming MosFET (QDIM) with the drain to the nDIM pin and the source to GND. Apply an external logic-level PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn off QDIM if no signal is present. 2. Connect the anode of a Schottky diode (DDIM) to the nDIM pin. Apply an external inverted logic-level PWM signal to the cathode of the same diode. A minimum on-time must be maintained in order for PWM dimming to operate in the linear region of its transfer function. Because the controller is disabled during dimming, the PWM pulse must be long enough such that the energy intercepted from the input is greater than or equal to the energy being put into the LEDs. For boost and buck-boost regulators, the following condition must be maintained: 20 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Feature Description (continued) 2 x ILED x VO X L1 tPULSE = VIN2 (26) In the previous equation, tPULSE is the length of the PWM pulse in seconds. 7.3.11 Startup Regulator (VCC LDO) The LM3429 includes a high voltage, low dropout (LDO) bias regulator. When power is applied, the regulator is enabled and sources current into an external capacitor connected to the VCC pin. The VCC output voltage is 6.9V nominally and the supply is internally current limited to 20 mA (minimum). The recommended bypass capacitance range for the VCC regulator is 2.2 F to 3.3 F. The output of the VCC regulator is monitored by an internal UVLO circuit that protects the device during startup, normal operation, and shutdown from attempting to operate with insufficient supply voltage. 7.3.12 Thermal Shutdown The LM3429 includes thermal shutdown. If the die temperature reaches approximately 165C the device will shut down (GATE pin low), until it reaches approximately 140C where it turns on again. 7.4 Device Functional Modes This device has no functional modes. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 21 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI's customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality. 8.1 Application Information 8.1.1 Inductor The inductor (L1) is the main energy storage device in a switching regulator. Depending on the topology, energy is stored in the inductor and transfered to the load in different ways (as an example, buck-boost operation is detailed in the Current Regulators section). The size of the inductor, the voltage across it, and the length of the switching subinterval (tON or tOFF) determines the inductor current ripple (iL-PP). In the design process, L1 is chosen to provide a desired iL-PP. For a buck regulator the inductor has a direct connection to the load, which is good for a current regulator. This requires little to no output capacitance therefore iL-PP is basically equal to the LED ripple current iLED-PP. However, for boost and buck-boost regulators, there is always an output capacitor which reduces iLED-PP, therefore the inductor ripple can be larger than in the buck regulator case where output capacitance is minimal or completely absent. In general, iLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED). Therefore, for the buck regulator with no output capacitance, iL-PP should also be less than 40% of ILED. For the boost and buck-boost topologies, iL-PP can be much higher depending on the output capacitance value. However, iL-PP is suggested to be less than 100% of the average inductor current (IL) to limit the RMS inductor current. L1 is also suggested to have an RMS current rating at least 25% higher than the calculated minimum allowable RMS inductor current (IL-RMS). 8.1.2 LED Dynamic Resistance (rD) When the load is a string of LEDs, the output load resistance is the LED string dynamic resistance plus RSNS. LEDs are PN junction diodes, and their dynamic resistance shifts as their forward current changes. Dividing the forward voltage of a single LED (VLED) by the forward current (ILED) leads to an incorrect calculation of the dynamic resistance of a single LED (rLED). The result can be 5 to 10 times higher than the true rLED value. Figure 25. Dynamic Resistance Obtaining rLED is accomplished by referring to the manufacturer's LED I-V characteristic. It can be calculated as the slope at the nominal operating point as shown in Figure 25. For any application with more than 2 series LEDs, RSNS can be neglected allowing rD to be approximated as the number of LEDs multiplied by rLED. 22 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Application Information (continued) 8.1.3 Output Capacitor For boost and buck-boost regulators, the output capacitor (CO) provides energy to the load when the recirculating diode (D1) is reverse biased during the first switching subinterval. An output capacitor in a buck topology will simply reduce the LED current ripple (iLED-PP) below the inductor current ripple (iL-PP). In all cases, CO is sized to provide a desired iLED-PP. As mentioned in the Inductor section, iLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED). CO should be carefully chosen to account for derating due to temperature and operating voltage. It must also have the necessary RMS current rating. Ceramic capacitors are the best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R dieletric rating is suggested. 8.1.4 Input Capacitors The input capacitance (CIN) provides energy during the discontinuous portions of the switching period. For buck and buck-boost regulators, CIN provides energy during tON and during tOFF, the input voltage source charges up CIN with the average input current (IIN). For boost regulators, CIN only needs to provide the ripple current due to the direct connection to the inductor. CIN is selected given the maximum input voltage ripple (vIN-PP) which can be tolerated. vIN-PP is suggested to be less than 10% of the input voltage (VIN). An input capacitance at least 100% greater than the calculated CIN value is recommended to account for derating due to temperature and operating voltage. When PWM dimming, even more capacitance can be helpful to minimize the large current draw from the input voltage source during the rising transition of the LED current waveform. The chosen input capacitors must also have the necessary RMS current rating. Ceramic capacitors are again the best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R dieletric rating is suggested. For most applications, TI recommends bypassing the VIN pin with an 0.1-F ceramic capacitor placed as close as possible to the pin. In situations where the bulk input capacitance may be far from the LM3429 device, a 10- series resistor can be placed between the bulk input capacitance and the bypass capacitor, creating a 150 kHz filter to eliminate undesired high frequency noise coupling. 8.1.5 N-Channel MosFET (NFET) The LM3429 requires an external NFET (Q1) as the main power MosFET for the switching regulator. Q1 is recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe operation during the ringing of the switch node. In practice, all switching regulators have some ringing at the switch node due to the diode parasitic capacitance and the lead inductance. The current rating is recommended to be at least 10% higher than the average transistor current. The power rating is then verified by calculating the power loss given the RMS transistor current and the NFET on-resistance (RDS-ON). In general, the NFET should be chosen to minimize total gate charge (Qg) whenever switching frequencies are high and minimize RDS-ON otherwise. This will minimize the dominant power losses in the system. Frequently, higher current NFETs in larger packages are chosen for better thermal performance. 8.1.6 Re-Circulating Diode A re-circulating diode (D1) is required to carry the inductor current during tOFF. The most efficient choice for D1 is a Schottky diode due to low forward voltage drop and near-zero reverse recovery time. Similar to Q1, D1 is recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe operation during the ringing of the switch node and a current rating at least 10% higher than the average diode current. The power rating is verified by calculating the power loss through the diode. This is accomplished by checking the typical diode forward voltage from the I-V curve on the product data sheet and multiplying by the average diode current. In general, higher current diodes have a lower forward voltage and come in better performing packages minimizing both power losses and temperature rise. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 23 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8.2 Typical Applications 8.2.1 Basic Topology Schematics L1 D1 VIN CIN RT 1 CCMP 2 LM3429 VIN HSN HSP COMP 14 13 RHSN CFS RHSP RSNS CO RFS RCSH 3 CT 4 CSH IS RCT VCC ILED 12 11 CBYP 5 6 RUV2 PWM GATE OVP PGND 10 Q1 9 RLIM ROV2 DAP RUVH RUV1 AGND 7 nDIM NC 8 COV Q3 ROV1 Figure 26. Boost Regulator (VIN < VO) 24 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) VIN CIN 1 LM3429 VIN HSN 14 RT 2 HSP COMP 13 RHSN CFS RHSP RSNS CO CCMP RFS 3 CSH IS 12 D1 ILED L1 RCSH 4 RCT VCC 11 CT CBYP 5 AGND GATE 10 Q1 ROV2 6 RUV2 OVP PGND 7 PWM RLIM Q2 DAP RUVH RUV1 9 nDIM NC 8 COV Q3 ROV1 Figure 27. Buck Regulator (VIN > VO) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 25 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) L1 D1 VIN CIN ILED RT 1 CCMP 2 VIN LM3429 HSN HSP COMP 14 13 RHSN CO CFS RHSP RSNS VIN RFS RCSH 3 CT 4 CSH IS RCT VCC 12 11 CBYP 5 AGND GATE 10 Q1 ROV2 6 RUV2 OVP PGND 7 PWM RLIM VIN DAP RUVH RUV1 9 nDIM NC Q2 8 COV Q3 ROV1 Figure 28. Buck-Boost Regulator 8.2.1.1 Design Requirements Number of series LEDs: N Single LED forward voltage: VLED Single LED dynamic resistance: rLED Nominal input voltage: VIN Input voltage range: VIN-MAX, VIN-MIN Switching frequency: fSW Current sense voltage: VSNS Average LED current: ILED Inductor current ripple: iL-PP LED current ripple: iLED-PP Peak current limit: ILIM Input voltage ripple: vIN-PP Output OVLO characteristics: VTURN-OFF, VHYSO Input UVLO characteristics: VTURN-ON, VHYS 26 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) 8.2.1.2 Detailed Design Procedure 8.2.1.2.1 Operating Point Given the number of series LEDs (N), the forward voltage (VLED) and dynamic resistance (rLED) for a single LED, solve for the nominal output voltage (VO) and the nominal LED string dynamic resistance (rD): VO = N x VLED (27) rD = N x rLED (28) Solve for the ideal nominal duty cycle (D): Buck D= VO VIN (29) VO - VIN VO (30) Boost D= Buck-boost D= VO VO + VIN (31) Using the same equations, find the minimum duty cycle (DMIN) using maximum input voltage (VIN-MAX) and the maximum duty cycle (DMAX) using the minimum input voltage (VIN-MIN). Also, remember that D' = 1 - D. 8.2.1.2.2 Switching Frequency Set the switching frequency (fSW) by assuming a CT value of 1 nF and solving for RT: Buck (Constant Ripple vs. VIN) RT = 25 x ( VIN - VO ) fSW x CT X VIN (32) Buck (Constant Ripple vs. VO) 2 RT = 25 x (VIN x VO - VO fSW x C T x ) 2 VIN (33) Boost and Buck-Boost 25 RT = fSW x C T (34) 8.2.1.2.3 Average LED Current For all topologies, set the average LED current (ILED) knowing the desired current sense voltage (VSNS) and solving for RSNS: VSNS RSNS = ILED (35) If the calculated RSNS is too far from a desired standard value, then VSNS must be adjusted to obtain a standard value. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 27 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) Setup the suggested signal current of 100 A by assuming RCSH = 12.4 k and solving for RHSP: ILED x RCSH x RSNS RHSP = 1.24V (36) If the calculated RHSP is too far from a desired standard value, then RCSH can be adjusted to obtain a standard value. 8.2.1.2.4 Inductor Ripple Current Set the nominal inductor ripple current (iL-PP) by solving for the appropriate inductor (L1): Buck L1 (VIN VO ) x D 'iL PP x fSW (37) Boost and Buck-Boost VIN x D L1 'iL PP x fSW (38) To set the worst case inductor ripple current, use VIN-MAX and DMIN when solving for L1. The minimum allowable inductor RMS current rating (IL-RMS) can be calculated as: Buck IL-RMS = ILED x 1 'IL-PP* x 1+ 12 ILED (c) 2 (39) Boost and Buck-Boost 1 'IL-PP x D' * x x 1+ IL-RMS = 12 ILED D' ILED (c) 2 (40) 8.2.1.2.5 LED Ripple Current Set the nominal LED ripple current (iLED-PP), by solving for the output capacitance (CO): Buck CO = 'iL - PP 8 x fSW x rD x 'iLED - PP (41) Boost and Buck-Boost ILED x D Co rD x 'iLED PP (42) To set the worst case LED ripple current, use DMAX when solving for CO. The minimum allowable RMS output capacitor current rating (ICO-RMS) can be approximated: Buck ICO - RMS = 28 uiLED - PP 12 Submit Documentation Feedback (43) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) Boost and Buck-boost ICO-RMS = ILED x DMAX 1-DMAX (44) 8.2.1.2.6 Peak Current Limit Set the peak current limit (ILIM) by solving for the transistor path sense resistor (RLIM): R LIM = 245 mV ILIM (45) 8.2.1.2.7 Loop Compensation Using a simple first order peak current mode control model, neglecting any output capacitor ESR dynamics, the necessary loop compensation can be determined. First, the uncompensated loop gain (TU) of the regulator can be approximated: Buck TU = TU0 x 1 s * 1+ ZP1 (c) (46) Boost and Buck-Boost s * 1 ZZ1 (c) TU = TU0 x s * 1+ ZP1 (c) (47) Where the pole (P1) is approximated: 3 Buck 1 rD x CO ZP1 = (48) 3 Boost 2 rD x CO ZP1 = (49) 3 Buck-Boost ZP1 = 1+ D rD x CO (50) And the RHP zero (Z1) is approximated: Boost ZZ1 = rD x Dc2 L1 (51) Buck-Boost ZZ1 = rD x Dc2 D x L1 (52) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 29 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) And the uncompensated DC loop gain (TU0) is approximated: Buck 500V x RCSH x RSNS 620V = RHSP x R LIM ILED x RLIM (53) Dc x 500V x RCSH x RSNS Dc x 310V = 2 x RHSP x R LIM ILED x R LIM (54) TU0 = Boost TU0 = Buck-Boost Dc x 500V x RCSH x RSNS Dc x 620V TU0 = = (1+ D) x RHSP x R LIM (1+ D) x ILED x R LIM (55) For all topologies, the primary method of compensation is to place a low-frequency dominant pole (P2) which will ensure that there is ample phase margin at the crossover frequency. This is accomplished by placing a capacitor (CCMP) from the COMP pin to GND, which is calculated according to the lower value of the pole and the RHP zero of the system (shown as a minimizing function): ZP2 = min(Z P1, ZZ1) 5 x TU0 (56) 1 C CMP = P2 x5x106 (57) If analog dimming is used, CCMP should be approximately 4x larger to maintain stability as the LEDs are dimmed to zero. A high frequency compensation pole (P3) can be used to attenuate switching noise and provide better gain margin. Assuming RFS = 10 , CFS is calculated according to the higher value of the pole and the RHP zero of the system (shown as a maximizing function): ZP3 = max (ZP1, ZZ1) x 10 CFS = (58) 1 10 x ZP3 (59) The total system loop gain (T) can then be written as: Buck T = TU0 x 1 s * s * 1+ ZP2 x 1+ ZP3 (c) (c) (60) s * 1 ZZ1 (c) T = TU0 x s * s * s * 1+ ZP1 x 1+ ZP2 x 1+ ZP3 (c) (c) (c) (61) s * 1+ ZP1 x (c) Boost and Buck-boost 8.2.1.2.8 Input Capacitance Set the nominal input voltage ripple (vIN-PP) by solving for the required capacitance (CIN): Buck 30 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) CIN = ILED x (1 - D) x D 'VIN-PP x fSW (62) Boost CIN = 'iL-PP 8 x 'VIN-PP x fSW (63) Buck-Boost CIN = ILED x D 'VIN-PP x fSW (64) Use DMAX to set the worst case input voltage ripple, when solving for CIN in a buck-boost regulator and DMID = 0.5 when solving for CIN in a buck regulator. The minimum allowable RMS input current rating (ICIN-RMS) can be approximated: Buck ICIN - RMS = ILED x DMID x (1-DMID) (65) Boost ICIN-RMS = 'iL-PP 12 (66) Buck-Boost ICIN-RMS = ILED x DMAX 1-DMAX (67) 8.2.1.2.9 NFET The NFET voltage rating should be at least 15% higher than the maximum NFET drain-to-source voltage (VTMAX): Buck VT - MAX = VIN - MAX (68) VT - MAX = VO (69) Boost Buck-Boost VT - MAX = VIN - MAX + VO (70) The current rating should be at least 10% higher than the maximum average NFET current (IT-MAX): Buck IT-MAX = DMAX x ILED (71) Boost and Buck-Boost DMAX IT-MAX = xI 1 - DMAX LED (72) Approximate the nominal RMS transistor current (IT-RMS) : Buck Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 31 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) IT- RMS = ILED x D (73) Boost and Buck-Boost ILED x D IT - RMS = Dc (74) Given an NFET with on-resistance (RDS-ON), solve for the nominal power dissipation (PT): 2 PT = IT - RMS x R DSON (75) 8.2.1.2.10 Diode The Schottky diode voltage rating should be at least 15% higher than the maximum blocking voltage (VRD-MAX): Buck VRD-MAX = VIN-MAX (76) Boost VRD-MAX = VO (77) Buck-Boost VRD-MAX = VIN-MAX + VO (78) The current rating should be at least 10% higher than the maximum average diode current (ID-MAX): Buck ID-MAX = (1 - DMIN) x ILED (79) Boost and Buck-Boost ID-MAX = ILED (80) Replace DMAX with D in the ID-MAX equation to solve for the average diode current (ID). Given a diode with forward voltage (VFD), solve for the nominal power dissipation (PD): PD = ID x VFD (81) 8.2.1.2.11 Output OVLO For boost and buck-boost regulators, output OVLO is programmed with the turn-off threshold voltage (VTURN-OFF) and the desired hysteresis (VHYSO). To set VHYSO, solve for ROV2: VHYSO ROV2 = 20 PA (82) To set VTURN-OFF, solve for ROV1: Boost ROV1 = 1.24V x ROV2 VTURN - OFF - 1.24V (83) Buck-Boost R OV1 = 32 1.24V x R OV2 VTURN - OFF - 620 mV Submit Documentation Feedback (84) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) A small filter capacitor (COVP = 47 pF) should be added from the OVP pin to ground to reduce coupled switching noise. 8.2.1.2.12 Input UVLO For all topologies, input UVLO is programmed with the turn-on threshold voltage (VTURN-ON) and the desired hysteresis (VHYS). Method #1: If no PWM dimming is required, a two resistor network can be used. To set VHYS, solve for RUV2: VHYS RUV2 = 20 PA (85) To set VTURN-ON, solve for RUV1: RUV1 = 1.24V x RUV2 VTURN - ON - 1.24V (86) Method #2: If PWM dimming is required, a three resistor network is suggested. To set VTURN-ON, assume RUV2 = 10 k and solve for RUV1 as in Method #1. To set VHYS, solve for RUVH: RUVH = R UV1 x (VHYS - 20 PA x RUV2) 20 PA x (RUV1 + R UV2) (87) 8.2.1.2.13 PWM Dimming Method PWM dimming can be performed several ways: Method #1: Connect the dimming MosFET (Q3) with the drain to the nDIM pin and the source to GND. Apply an external PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn off Q3. Method #2: Connect the anode of a Schottky diode to the nDIM pin. Apply an external inverted PWM signal to the cathode of the same diode. 8.2.1.2.14 Analog Dimming Method Analog dimming can be performed several ways: Method #1: Place a potentiometer in series with the RCSH resistor to dim the LED current from the nominal ILED to near zero. Method #2: Connect a controlled current source as detailed in the Analog Dimming section to the CSH pin. Increasing the current sourced into the CSH node will decrease the LEDs from the nominal ILED to zero current. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 33 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) 8.2.2 Buck-Boost Application - 6 LEDs at 1 A 10V 70V VIN L1 CIN D1 RT 1 CCMP RCSH CT 2 3 4 VIN LM3429 HSN HSP COMP CSH IS RCT VCC 14 RHSN 13 RHSP 1A ILED CO 12 CFS 11 VIN CBYP 5 AGND GATE RSNS RFS 10 Q1 ROV2 6 RUV2 OVP PGND 9 RLIM VIN DAP 7 nDIM NC Q2 8 COV RUV1 ROV1 Figure 29. Buck-Boost Application - 6 LEDs at 1 A Schematic 8.2.2.1 Design Requirements N=6 VLED = 3.5 V rLED = 325 m VIN = 24 V VIN-MIN = 10 V VIN-MAX = 70 V fSW = 700 kHz VSNS = 100 mV ILED = 1A iL-PP = 500 mA iLED-PP = 50 mA vIN-PP = 100 mV ILIM = 6A VTURN-ON = 10 V VHYS = 3 V 34 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) VTURN-OFF = 40 V VHYSO = 10 V 8.2.2.2 Detailed Design Procedure 8.2.2.2.1 Operating Point Solve for VO and rD: VO = N x VLED = 6 x 3.5V = 21V (88) rD = N x rLED = 6 x 325 m: = 1. 95: (89) Solve for D, D', DMAX, and DMIN: D= VO 21V = = 0.467 VO + VIN 21V + 24V (90) D' = 1 - D = 1 - 0. 467 = 0. 533 DMIN = DMAX = (91) VO 21V = = 0.231 VO + VIN-MAX 21V + 70V (92) VO 21V = = 0.677 VO + VIN-MIN 21V + 10V (93) 8.2.2.2.2 Switching Frequency Assume CT = 1 nF and solve for RT: RT = 25 25 = = 35.7 k: fSW x CT 700 kHz x 1 nF (94) The closest standard resistor is actually 35.7 k therefore the fSW is: fSW = 25 25 = = 700 kHz RT x CT 35.7 k: x 1 nF (95) The chosen components from step 2 are: CT = 1 nF RT = 35.7 k: (96) 8.2.2.2.3 Average LED Current Solve for RSNS: V 100 mV RSNS = SNS = = 0.1: ILED 1A (97) Assume RCSH = 12.4 k and solve for RHSP: ILED x RCSH x RSNS 1A x 12.4 k : x 0.1: RHSP = = = 1.0 k: 1.24V 1.24V (98) The closest standard resistor for RSNS is actually 0.1 and for RHSP is actually 1 k therefore ILED is: 1.24V x RHSP 1.24V x 1.0 k: ILED = = = 1.0A R SNS x R CSH 0.1: x 12.4 k: (99) The chosen components from step 3 are: Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 35 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) RS NS = 0.1: R CSH = 12.4 k : RHSP = RHSN = 1 k: (100) 8.2.2.2.4 Inductor Ripple Current Solve for L1: L1 = VIN x D 24V x 0. 467 = = 32 PH 'iL- PP x fSW 500 mA x 700 kHz (101) The closest standard inductor is 33 H therefore the actual iL-PP is: 'iL- PP = VIN x D 24V x 0. 467 = 485 mA = L1 x fSW 33 PH x 700 kHz (102) Determine minimum allowable RMS current rating: 2 I 1 x 'iL - PP x Dc* IL - RMS = LED x 1+ 12 (c) ILED Dc 2 1 x 485 mA x 0.533* 1A x 1+ 12 (c) 1A 0. 533 IL - RMS = 1.88A IL - RMS = (103) The chosen component from step 4 is: L1 = 33 PH (104) 8.2.2.2.5 Output Capacitance Solve for CO: CO = CO = ILED x D rD x 'iLED- PP x fSW 1A x 0. 467 = 6.84 PF 1.95: x 50 mA x 7 00 kHz (105) The closest standard capacitor is 6.8 F therefore the actual iLED-PP is: I xD 'iLED- PP = LED rD x CO x fSW 'iLED- PP = 1A x 0. 467 = 50 mA 1.95 : x 6.8 PF x 7 00 kHz (106) Determine minimum allowable RMS current rating: ICO- RMS = ILED x DMAX 0.677 = 1.45A = 1A x 1- DMAX 1- 0.677 (107) The chosen components from step 5 are: CO = 6.8 PF 36 Submit Documentation Feedback (108) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) 8.2.2.2.6 Peak Current Limit Solve for RLIM: RLIM = 245 mV 245 mV = = 0.041: ILIM 6A (109) The closest standard resistor is 0.04 therefore ILIM is: ILIM = 245 mV 245 mV = = 6.13A RLIM 0.04 : (110) The chosen component from step 6 is: RLIM = 0.04: (111) 8.2.2.2.7 Loop Compensation P1 is approximated: ZP1 = 1.467 1+D rad = = 110k sec rD x CO 1.95: x 6.8 PF (112) Z1 is approximated: rD x Dc2 1.95: x 0.5332 rad ZZ1 = = = 37k D x L1 0.467 x 33 PH sec (113) TU0 is approximated: 0.533 x 620V Dc x 620V TU0 = = = 5630 1 . 467 x 1A x 0.04: (1+ D) x ILED x R LIM (114) To ensure stability, calculate P2: rad 37k min(ZP1, ZZ1) ZZ1 sec rad ZP2 = = = = 1.173 5 x 5630 5 x 5630 5 x TU0 sec (115) Solve for CCMP: CCMP = 1 1 = = 0.17 F P2x 5x106 1.173 rad x 5x10 6 sec (116) To attenuate switching noise, calculate P3: ZP3 = max ZP1, ZZ1 x 10 = ZP1 x 10 ZP3 = 110 k rad rad x 10 = 1.1M sec sec (117) Assume RFS = 10 and solve for CFS: CFS = 1 = 10: x ZP3 1 10: x 1.1M rad sec = 0.091 PF (118) The chosen components from step 7 are: CCOMP = 0.22 PF RFS = 10: CFS = 0.1 PF (119) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 37 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) 8.2.2.2.8 Input Capacitance Solve for the minimum CIN: CIN = ILED x D 1A x 0. 467 = = 6.66 PF 'vIN- PP x fSW 100 mV x 700 kHz (120) To minimize power supply interaction a 200% larger capacitance of approximately 14 F is used, therefore the actual vIN-PP is much lower. Because high voltage ceramic capacitor selection is limited, three 4.7 F X7R capacitors are chosen. Determine minimum allowable RMS current rating: IIN- RMS = ILED x DMAX 0.677 = 1.45A = 1A x 1- DMAX 1- 0.677 (121) The chosen components from step 8 are: CIN = 3 x 4.7 PF (122) 8.2.2.2.9 NFET Determine minimum Q1 voltage rating and current rating: VT - MAX = VIN - MAX + VO = 70V + 21V = 91V IT- MAX = (123) 0. 677 x 1A = 2.1A 1- 0.677 (124) A 100-V NFET is chosen with a current rating of 32A due to the low RDS-ON = 50 m. Determine IT-RMS and PT: IT - RMS = PT = ILED 1A x D= x 0.467 = 1. 28A 0. 533 Dc 2 IT- RMS (125) 2 x RDSON = 1. 28A x 50 m: = 82 mW (126) The chosen component from step 9 is: Q1 o 32A, 100V, DPAK (127) 8.2.2.2.10 Diode Determine minimum D1 voltage rating and current rating: VRD - MAX = VIN - MAX + VO = 70V + 21V = 91V (128) ID - MAX = ILED = 1A (129) A 100-V diode is chosen with a current rating of 12 A and VDF = 600 mV. Determine PD: PD = ID x VFD = 1A x 600 mV = 600 mW (130) The chosen component from step 10 is: D1 o 12A, 100V, DPAK (131) 8.2.2.2.11 Input UVLO Solve for RUV2: 38 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Typical Applications (continued) R UV2 = VHYS 3V = = 150 k: 20 P A 20 PA (132) The closest standard resistor is 150 k therefore VHYS is: VHYS = RUV2 x 20 P A = 150 k: x 20 P A = 3V (133) Solve for RUV1: 1.24V x R UV2 1.24V x 150 k: R UV1 = = = 21.2 k: VTURN - ON - 1.24V 10V -1.24V (134) The closest standard resistor is 21 k making VTURN-ON: VTURN - ON = 1.24V x (R UV1 + R UV2) R UV1 VTURN- ON = 1.24V x (21 k: + 150 k:) = 10.1V 21 k: (135) The chosen components from step 11 are: RUV1 = 21 k: RUV2 = 150 k: (136) 8.2.2.2.12 Output OVLO Solve for ROV2: ROV2 = VHYSO 10V = = 500 k: 20 P A 20 P A (137) The closest standard resistor is 499 k therefore VHYSO is: VHYSO = ROV2 x 20 PA = 499 k: x 20 PA = 9.98V (138) Solve for ROV1: 1.24V x ROV2 1.24V x 499 k: R OV1 = = = 15.7 k: VTURN - OFF - 0.62V 40V - 0.62V (139) The closest standard resistor is 15.8 k making VTURN-OFF: VTURN-OFF = VTURN-OFF = 1.24V x (0.5 x ROV1 + ROV2) ROV1 1.24V x (0.5 x 15.8 k: + 499 k:) = 39.8V 15.8 k: (140) The chosen components from step 12 are: ROV1 = 15.8 k: ROV2 = 499 k: (141) Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 39 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Typical Applications (continued) Table 1. Design 1 Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 0.22 F X7R 10% 25 V MURATA GRM21BR71E224KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.1 F X7R 10% 25 V MURATA GRM21BR71E104KA01L 3 CIN 4.7 F X7R 10% 100 V TDK C5750X7R2A475K 1 CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF 1 L1 33 H 20% 6.3 A COILCRAFT MSS1278-333MLB 1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682 1 Q2 PNP 150 V 600 m A FAIRCHILD MMBT5401 1 RCSH 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1 k 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.04 1% 1W VISHAY WSL2512R0400FEA 1 ROV1 15.8 k 1% VISHAY CRCW080515K8FKEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.1 1% 1W VISHAY WSL2512R1000FEA 1 RT 35.7 k 1% VISHAY CRCW080535K7FKEA 1 RUV1 21 k 1% VISHAY CRCW080521K0FKEA 1 RUV2 150 k 1% VISHAY CRCW0805150KFKEA 8.2.2.3 Application Curve 100 EFFICIENCY (%) 95 90 85 80 75 70 0 16 32 48 VIN (V) 64 80 Figure 30. Buck-Boost Efficiency vs Input Voltage, VO= 6 LEDs 40 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 8.2.3 Boost PWM Dimming Application - 9 LEDs at 1 A 8V - 28V VIN L1 CIN D1 RT 1 CCMP 2 LM3429 VIN HSN HSP COMP 14 RHSN 13 RHSP CFS RSNS RFS RCSH 3 CT 4 CSH IS RCT VCC 11 1A ILED CBYP 5 6 RUV2 PWM GATE OVP PGND 10 Q1 9 RLIM ROV2 DAP RUVH RUV1 AGND CO 12 7 nDIM NC 8 COV Q2 ROV1 Figure 31. Boost PWM Dimming Application - 9 LEDs at 1 A Schematic 8.2.3.1 Detailed Design Procedure Table 2. Design 2 Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 2 CCMP, CFS 0.1 F X7R 10% 25 V MURATA GRM21BR71E104KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 2, 1 CIN, CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60 V 5 A COMCHIP CDBC560-G 1 L1 33 H 20% 6.3 A COILCRAFT MSS1278-333MLB 1 Q1 NMOS 60 V 8 A VISHAY SI4436DY 1 Q2 NMOS 60 V 115 mA ON SEMI 2N7002ET1G 2 RCSH, ROV1 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1 k 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.06 1% 1 W VISHAY WSL2512R0600FEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.1 1% 1 W VISHAY WSL2512R1000FEA 1 RT 35.7 k 1% VISHAY CRCW080535K7FKEA 1 RUV1 1.82 k 1% VISHAY CRCW08051K82FKEA 1 RUV2 10 k 1% VISHAY CRCW080510KFKEA 1 RUVH 17.8 k 1% VISHAY CRCW080517K8FKEA Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 41 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8.2.4 Buck-Boost Analog Dimming Application - 4 LEDs at 2A 10V 30V VIN L1 D1 CIN RT 1 LM3429 VIN HSN 14 RHSN 13 RHSP 2A ILED CO CCMP RCSH 2 HSP COMP CFS 3 CSH IS RSNS 12 VIN RADJ RFS CT 4 RCT VCC 11 CBYP 5 AGND GATE 10 Q1 ROV2 RUV2 6 OVP PGND 9 RLIM VIN DAP 7 nDIM NC Q2 8 COV RUV1 ROV1 Figure 32. Buck-Boost Analog Dimming Application - 4 LEDs at 2 A Schematic 8.2.4.1 Detailed Design Procedure Table 3. Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 1 F X7R 10% 10 V MURATA GRM21BR71A105KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.1 F X7R 10% 50 V MURATA GRM21BR71E104KA01L 2, 1 CIN, CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60 V 5 A VISHAY CDBC560-G 1 L1 22 H 20% 7.2 A COILCRAFT MSS1278-223MLB 1 Q1 NMOS 60 V 8 A VISHAY SI4436DY 1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401 1 RADJ 1-M potentiometer BOURNS 3352P-1-105 1 RCSH 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1 k 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.04 1% 1 W VISHAY WSL2512R0400FEA 1 ROV1 18.2 k 1% VISHAY CRCW080518K2FKEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.05 1% 1 W VISHAY WSL2512R0500FEA 1 RT 41.2 k 1% VISHAY CRCW080541K2FKEA 42 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Table 3. Bill of Materials (continued) QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 RUV1 21 k 1% VISHAY CRCW080521K0FKEA 1 RUV2 150 k 1% VISHAY CRCW0805150KFKEA Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 43 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8.2.5 Boost Analog Dimming Application - 12 LEDs at 700 mA L1 18V - 38V VIN D1 CIN RT 1 CCMP 2 VREF VIN LM3429 HSN HSP COMP 14 RHSN 13 RHSP CFS RFS Q4 Q3 3 RMAX RBIAS CSH IS RCT VCC 12 CO CT Q2 RADJ RSNS 4 11 700 mA ILED CBYP RCSH 5 VIN 6 RUV2 AGND GATE OVP PGND 10 Q1 9 RLIM ROV2 DAP 7 nDIM NC 8 COV RUV1 ROV1 Figure 33. Boost Analog Dimming Application - 12 LEDs at 700 mA Schematic 8.2.5.1 Detailed Design Procedure Table 4. Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 1 F X7R 10% 10 V MURATA GRM21BR71A105KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.1 F X7R 10% 50 V MURATA GRM21BR71E104KA01L 2, 1 CIN, CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF 1 L1 47 H 20% 5.3 A COILCRAFT MSS1278-473MLB 1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682 1 Q2 NPN 40 V 200 mA FAIRCHILD MMBT3904 1 Q3, Q4 (dual pack) Dual PNP 40 V 200 mA FAIRCHILD FFB3906 1 RADJ 100 k potentiometer BOURNS 3352P-1-104 1 RBIAS 40.2 k 1% VISHAY CRCW080540K2FKEA 1 RCSH, ROV1, RUV1 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1.05 k 1% VISHAY CRCW08051K05FKEA 1 RLIM 0.06 1% 1 W VISHAY WSL2512R0600FEA 1 RMAX 4.99 k 1% VISHAY CRCW08054K99FKEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.15 1% 1 W VISHAY WSL2512R1500FEA 44 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Table 4. Bill of Materials (continued) QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 RT 35.7 k 1% VISHAY CRCW080535K7FKEA 1 RUV2 100 k 1% VISHAY CRCW0805100KFKEA 1 VREF 5 V precision reference TI LM4040 Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 45 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8.2.6 Buck-Boost PWM Dimming Application - 6 LEDs at 500 mA 10V 70V VIN L1 CIN D1 RT 1 CCMP 2 RCSH 3 CT 4 LM3429 VIN HSN HSP COMP CSH IS RCT VCC 14 RHSN 13 RHSP 500 mA ILED 12 CFS 11 AGND GATE 10 RSNS VIN CBYP 5 CO RFS Q1 ROV2 6 RUV2 OVP PGND 9 VIN DAP RUVH 7 nDIM NC Q2 8 D2 COV RUV1 ROV1 PWM Figure 34. Buck-Boost PWM Dimming Application - 6 LEDs at 500 mA 8.2.6.1 Detailed Design Procedure Table 5. Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 0.68 F X7R 10% 25 V MURATA GRM21BR71E684KA88L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.1 F X7R 10% 25 V MURATA GRM21BR71E104KA01L 3 CIN 4.7 F X7R 10% 100 V TDK C5750X7R2A475K 1 CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF 1 D2 Schottky 30 V 500 mA ON SEMI BAT54T1G 1 L1 68 H 20% 4.3 A COILCRAFT MSS1278-683MLB 1 Q1 NMOS 100 V 32 A VISHAY FDD3682 1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401 1 RCSH 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1 k 1% VISHAY CRCW08051K00FKEA 1 ROV1 15.8 k 1% VISHAY CRCW080515K8FKEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.2 1% 1 W VISHAY WSL2512R2000FEA 1 RT 35.7 k 1% VISHAY CRCW080535K7FKEA 1 RUV1 1.43 k 1% VISHAY CRCW08051K43FKEA 1 RUV2 10 k 1% VISHAY CRCW080510K0FKEA 46 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Table 5. Bill of Materials (continued) QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 RUVH 17.4 k 1% VISHAY CRCW080517K4FKEA Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 47 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8.2.7 Buck Application - 3 LEDS at 1.25 A 15V 50V VIN CIN 1 LM3429 VIN HSN 14 RHSN RT 2 HSP COMP CFS RHSP 13 RSNS CO CCMP RFS 3 CSH IS RCT VCC D1 12 1.25A ILED RCSH 4 11 CT L1 CBYP 5 AGND GATE 10 Q1 ROV2 6 RUV2 OVP PGND 9 RLIM Q2 DAP 7 nDIM NC 8 COV RUV1 ROV1 Figure 35. Buck Application - 3 LEDS at 1.25 A Schematic 8.2.7.1 Detailed Design Procedure Table 6. Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 0.015 F X7R 10% 50 V MURATA GRM21BR71H153KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.01 F X7R 10% 50 V MURATA GRM21BR71H103KA01L 2 CIN 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 CO 1 F X7R 10% 50 V TDK C4532X7R1H105K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60V 5 A COMCHIP CDBC560-G 1 L1 22 H 20% 7.3 A COILCRAFT MSS1278-223MLB 1 Q1 NMOS 60 V 8 A VISHAY SI4436DY 1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401 1 RCSH 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RT 49.9 k 1% VISHAY CRCW080549K9FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1 k 1% VISHAY CRCW08051K00FKEA 1 RLIM 0.04 1% 1 W VISHAY WSL2512R0400FEA 1 ROV1 21.5 k 1% VISHAY CRCW080521K5FKEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.08 1% 1 W VISHAY WSL2512R0800FEA 48 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 Table 6. Bill of Materials (continued) QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 RUV1 11.5 k 1% VISHAY CRCW080511K5FKEA 1 RUV2 100 k 1% VISHAY CRCW0805100KFKEA Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 49 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 8.2.8 Buck-Boost Thermal Foldback Application - 8 LEDs at 2.5 A L1 15V 60V VIN VREF D1 CIN RT 1 RGAIN CCMP RNTC 2 VIN LM3429 HSN HSP COMP 14 RHSN 13 RHSP 2.5A ILED CO D2 RBIAS 3 RCSH CT 4 CSH IS RCT VCC 12 11 CBYP 5 VIN AGND GATE CFS 10 RSNS Q1 VIN RFS 6 RUV2 OVP PGND 9 ROV2 RLIM VIN DAP 7 nDIM NC Q2 8 COV RUV1 ROV1 Figure 36. Buck-Boost Thermal Foldback Application - 8 LEDs at 2.5 A Schematic 8.2.8.1 Detailed Design Procedure Table 7. Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 0.1 F X7R 10% 25 V MURATA GRM21BR71E104KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.1 F X7R 10% 25 V MURATA GRM21BR71E104KA01L 3 CIN 4.7 F X7R 10% 100 V TDK C5750X7R2A475K 1 CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF 1 L1 22 H 20% 7.2 A COILCRAFT MSS1278-223MLB 1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682 1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401 2 RCSH, ROV1 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 1 k 1% VISHAY CRCW08051K00FKEA 2 RLIM, RSNS 0.04 1% 1 W VISHAY WSL2512R0400FEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RT 49.9 k 1% VISHAY CRCW080549K9FKEA 1 RUV1 13.7 k 1% VISHAY CRCW080513K7FKEA 1 RUV2 150 k 1% VISHAY CRCW0805150KFKEA 50 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 8.2.9 SEPIC Application - 5 LEDs at 750 mA 9V - 36V VIN L1 CSEP D1 L2 CIN RT 1 CCMP 2 LM3429 VIN HSN HSP COMP 14 RHSN 13 RHSP CFS RSNS RFS RCSH CT 3 4 CSH IS RCT VCC 11 750 mA ILED CBYP 5 6 RUV2 AGND GATE OVP PGND CO 12 10 Q1 9 ROV2 DAP 7 nDIM NC 8 COV RUV1 ROV1 Figure 37. 5 LEDs at 750 mA 8.2.9.1 Detailed Design Procedure Table 8. Bill of Materials QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 LM3429 Boost controller TI LM3429MH 1 CCMP 0.47 F X7R 10% 25 V MURATA GRM21BR71E474KA01L 1 CF 2.2 F X7R 10% 16 V MURATA GRM21BR71C225KA12L 1 CFS 0.1 F X7R 10% 25 V MURATA GRM21BR71E104KA01L 2, 1 CIN, CO 6.8 F X7R 10% 50 V TDK C4532X7R1H685K 1 COV 47 pF COG/NPO 5% 50 V AVX 08055A470JAT2A 1 CSEP 1 F X7R 10% 100 V TDK C4532X7R2A105K 1 CT 1000 pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D 1 D1 Schottky 60 V 5 A COMCHIP CDBC560-G 1 L1, L2 68 H 20% 4.3 A COILCRAFT DO3340P-683 1 Q1 NMOS 60 V 8 A VISHAY SI4436DY 1 Q2 NMOS 60 V 115 mA ON SEMI 2N7002ET1G 1 RCSH 12.4 k 1% VISHAY CRCW080512K4FKEA 1 RFS 10 1% VISHAY CRCW080510R0FKEA 2 RHSP, RHSN 750 1% VISHAY CRCW0805750RFKEA 1 RLIM 0.04 1% 1 W VISHAY WSL2512R0400FEA 2 ROV1, RUV1 15.8 k 1% VISHAY CRCW080515K8FKEA 1 ROV2 499 k 1% VISHAY CRCW0805499KFKEA 1 RSNS 0.1 1% 1 W VISHAY WSL2512R1000FEA 1 RT 49.9 k 1% VISHAY CRCW080549K9FKEA Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 51 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com Table 8. Bill of Materials (continued) QTY PART ID PART VALUE MANUFACTURER PART NUMBER 1 RUV2 100 k 1% VISHAY CRCW0805100KFKEA 52 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 9 Power Supply Recommendations The device is designed to operate from an input voltage supply range from 4.5 V to 75 V. This input supply should be well regulated. If the input supply is located more than a few inches from the EVM or PCB, additional bulk capacitance may be required in addition to the ceramic bypass capacitors. 9.1 Input Supply Current Limit It is important to set the output current limit of your input supply to an appropriate value to avoid delays in your converter analysis and optimization. If not set high enough, current limit can be tripped during start-up or when your converter output power is increased, causing a foldback or shut-down condition. It is a common oversight when powering up a converter for the first time. 10 Layout 10.1 Layout Guidelines The performance of any switching regulator depends as much upon the layout of the PCB as the component selection. Following a few simple guidelines will maximimize noise rejection and minimize the generation of EMI within the circuit. Discontinuous currents are the most likely to generate EMI; therefore, take care when routing these paths. The main path for discontinuous current in the LM3429 buck regulator contains the input capacitor (CIN), the recirculating diode (D1), the N-channel MosFET (Q1), and the switch sense resistor (RLIM). In the LM3429 boost and buck-boost regulators, the discontinuous current flows through the output capacitor (CO), D1, Q1, and RLIM. In either case, this loop should be kept as small as possible and the connections between all the components should be short and thick to minimize parasitic inductance. In particular, the switch node (where L1, D1 and Q1 connect) should be just large enough to connect the components. To minimize excessive heating, large copper pours can be placed adjacent to the short current path of the switch node. The RCT, COMP, CSH, IS, HSP and HSN pins are all high-impedance inputs which couple external noise easily, therefore the loops containing these nodes should be minimized whenever possible. In some applications the LED or LED array can be far away (several inches or more) from the LM3429, or on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or separated from the rest of the regulator, the output capacitor should be placed close to the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 53 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 10.2 Layout Example Note critical paths and component placement: Minimize power loop containing discontinuous currents Minimize signal current loops (components close to IC) x Ground plane under IC for signal routing helps minimize noise coupling discontinuous switching frequency currents L1 D1 VIN Input Power CIN RT 1 GND VIN LM3429 HSN 14 RHSN 13 RHSP CCMP 2 HSP COMP CFS RSNS CO RFS RCSH 3 CSH IS RCT VCC 12 ILED CT 4 11 CBYP 5 RUV2 6 AGND GATE OVP PGND 7 PWM Q1 STAR GROUND RLIM 9 ROV2 DAP RUVH RUV1 10 nDIM NC 8 COV Q3 ROV1 Power Ground Figure 38. LM3429 Layout Guideline 54 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 LM3429, LM3429-Q1 www.ti.com SNVS616H - APRIL 2009 - REVISED JULY 2015 11 Device and Documentation Support 11.1 Device Support 11.1.1 Third-Party Products Disclaimer TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE. 11.2 Documentation Support 11.2.1 Related Documentation For related documentation see the following: * AN-1986 LM3429 Boost Evaluation Board, SNVA404 * AN-1985 LM3429 Buck-Boost Evaluation Board, SNVA403 11.3 Related Links The table below lists quick access links. Categories include technical documents, support and community resources, tools and software, and quick access to sample or buy. Table 9. Related Links PARTS PRODUCT FOLDER SAMPLE & BUY TECHNICAL DOCUMENTS TOOLS & SOFTWARE SUPPORT & COMMUNITY LM3429 Click here Click here Click here Click here Click here LM3429-Q1 Click here Click here Click here Click here Click here 11.4 Community Resources The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of Use. TI E2ETM Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help solve problems with fellow engineers. Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and contact information for technical support. The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of Use. TI E2ETM Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help solve problems with fellow engineers. Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and contact information for technical support. 11.5 Trademarks E2E is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 Submit Documentation Feedback 55 LM3429, LM3429-Q1 SNVS616H - APRIL 2009 - REVISED JULY 2015 www.ti.com 11.6 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 11.7 Glossary SLYZ022 -- TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. 56 Submit Documentation Feedback Copyright (c) 2009-2015, Texas Instruments Incorporated Product Folder Links: LM3429 LM3429-Q1 PACKAGE OPTION ADDENDUM www.ti.com 6-Feb-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (C) Device Marking (4/5) LM3429MH/NOPB ACTIVE HTSSOP PWP 14 94 Green (RoHS & no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429 MH LM3429MHX/NOPB ACTIVE HTSSOP PWP 14 2500 Green (RoHS & no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429 MH LM3429Q1MH/NOPB ACTIVE HTSSOP PWP 14 94 Green (RoHS & no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429 Q1MH LM3429Q1MHX/NOPB ACTIVE HTSSOP PWP 14 2500 Green (RoHS & no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM3429 Q1MH (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based flame retardants must also meet the <=1000ppm threshold requirement. (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish value exceeds the maximum column width. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 6-Feb-2020 continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. OTHER QUALIFIED VERSIONS OF LM3429, LM3429-Q1 : * Catalog: LM3429 * Automotive: LM3429-Q1 NOTE: Qualified Version Definitions: * Catalog - TI's standard catalog product * Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 6-Nov-2015 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) LM3429MHX/NOPB HTSSOP PWP 14 2500 330.0 12.4 LM3429Q1MHX/NOPB HTSSOP PWP 14 2500 330.0 12.4 Pack Materials-Page 1 B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant 6.95 5.6 1.6 8.0 12.0 Q1 6.95 5.6 1.6 8.0 12.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 6-Nov-2015 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM3429MHX/NOPB HTSSOP PWP 14 2500 367.0 367.0 35.0 LM3429Q1MHX/NOPB HTSSOP PWP 14 2500 367.0 367.0 35.0 Pack Materials-Page 2 MECHANICAL DATA PWP0014A MXA14A (Rev A) www.ti.com IMPORTANT NOTICE AND DISCLAIMER TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATASHEETS), DESIGN RESOURCES (INCLUDING REFERENCE DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES "AS IS" AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD PARTY INTELLECTUAL PROPERTY RIGHTS. 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