MAX767REAP+ - Maxim Integrated Products, Inc.

________________General Description

The MAX767 is a high-efficiency, synchronous buck

controller IC dedicated to converting a fixed 5V supply

into a tightly regulated 3.3V output. Two key features set

this device apart from similar, low-voltage step-down

switching regulators: high operating frequency and all

N-channel construction in the application circuit. The

300kHz operating frequency results in very small, low-

cost external surface-mount components.

The inductor, at 3.3µH for 5A, is physically at least five

times smaller than inductors found in competing solu-

tions. All N-channel construction and synchronous rectifi-

cation result in reduced cost and highest efficiency.

Efficiency exceeds 90% over a wide range of loading,

eliminating the need for heatsinking. Output capacitance

requirements are low, reducing board space and cost.

The MAX767 is a monolithic BiCMOS IC available in

20-pin SSOP packages. For other fixed output voltages

and package options, please consult the factory.

________________________Applications

Local 5V-to-3.3V DC-DC Conversion

Microprocessor Daughterboards

Power Supplies up to 10A or More

____________________________Features

♦>90% Efficiency

♦700µA Quiescent Supply Current

♦120µA Standby Supply Current

♦4.5V-to-5.5V Input Range

♦Low-Cost Application Circuit

♦All N-Channel Switches

♦Small External Components

♦Tiny Shrink-Small-Outline Package (SSOP)

♦Predesigned Applications:

Standard 5V to 3.3V DC-DC Converters up to 10A

High-Accuracy Pentium P54C VR-Spec Supply

♦Fixed Output Voltages Available:

3.3V (Standard)

3.45V (High-Speed Pentium™)

3.6V (PowerPC™)

_______________Ordering Information

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

________________________________________________________________ Maxim Integrated Products 1

BST

GND

TOP VIEW

MAX767

VCC

PGND

REF

GND

N.C.

GND

VCC

SYNC

SSOP

__________________Pin Configuration

OUTPUT

3.3V

AT 5A

INPUT

4.5V TO 5.5V

GND

MAX767

VCC

REF

3.3µH

BST

PGND

_________Typical Application Circuit

19-0224; Rev 3; 7/00

PART TEMP. RANGE PIN-

PACKAGE

MAX767CAP 0°C to +70°C 20 SSOP

MAX767RCAP 0°C to +70°C 20 SSOP

MAX767SCAP 0°C to +70°C 20 SSOP

Pentium is a trademark of Intel. PowerPC is a trademark of IBM.

MAX767C/D 0°C to +70°C Dice*

REF

TOL

±1.8%

—

Ordering Information continued at end of data sheet.

*Contact factory for dice specifications.

VOUT

(V)

3.3

3.45

3.6

—

MAX767TCAP 0°C to +70°C 20 SSOP ±1.2% 3.3

EVALUATION KIT MANUAL

FOLLOWS DATA SHEET

For price, delivery, and to place orders, please contact Maxim Distribution at 1-888-629-4642,

or visit Maxim’s website at www.maxim-ic.com.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional

operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to

absolute maximum rating conditions for extended periods may affect device reliability.

VCC to GND.................................................................-0.3V, +7V

PGND to GND ........................................................................±2V

BST to GND ...............................................................-0.3V, +15V

LX to BST.....................................................................-7V, +0.3V

Inputs/Outputs to GND

(ON, REF, SYNC, CS, FB, SS) .....................-0.3V, VCC + 0.3V

DL to PGND .....................................................-0.3V, VCC + 0.3V

DH to LX...........................................................-0.3V, BST + 0.3V

REF Short to GND.......................................................Momentary

REF Current.........................................................................20mA

Continuous Power Dissipation (TA= +70°C)

20-Pin SSOP (derate 8.00mW/°C above +70°C) ..........640mW

Operating Temperature Ranges:

MAX767CAP/MAX767_CAP.................................0°C to +70°C

MAX767EAP/MAX767_EAP ..............................-40°C to +85°C

Lead Temperature (soldering, 10s) .................................+300°C

PARAMETER

Oscillator Frequency

DH Sink/Source Current

MIN TYP MAX

260 300 340

UNITS

SS Source Current

2.50 4 6.5

(BST - LX) = 4.5V, DH = 2V

DL On Resistance

µA

Current-Limit Voltage

200

80 100 120 mV

VCC Fault Lockout Voltage

Ω

3.80 4.20

kHz

Line Regulation

High or low

0.1 %

Oscillator SYNC Range

SS Fault Sink Current

DH On Resistance

2mA

3.24 3.30 3.36

Ω

240 350 kHz

SYNC High Pulse Width

VCC Standby Current

High or low, (BST - LX) = 4.5V

120 200

200

µA

VCC Quiescent Current

0.7 1.0 mA

SYNC Low Pulse Width 200 ns

SYNC Rise/Fall Time

Output Voltage (FB)

200 ns

Oscillator Maximum Duty Cycle

VCC Input Supply Range

89 92

4.5 5.5 V

95 %

Input Low Voltage

3.17 3.35 3.46

0.8 V

Input High Voltage 2.40

3.32 3.50 3.60

VCC - 0.5 V

Input Current ±1 µA

DL Sink/Source Current 1 A

CONDITIONS

SYNC = 3.3V

SYNC = 0V or 5V

CS - FB

Falling edge, hysteresis = 1%

VCC = 4.5V to 5.5V

MAX767, MAX767R, MAX767S

ON = 0V, VCC = 5.5V

FB = CS = 3.5V

Not tested

SYNC = 3.3V

SYNC = 0V

SYNC, ON

0mV < (CS - FB) < 80mV,

4.5V < VCC < 5.5V

(includes load and

line regulation)

SYNC

SYNC, ON = 0V or 5V

DL = 2V

Load Regulation 2.5 %(CS - FB) = 0mV to 80mV

3.46 3.65 3.75

MAX767R

MAX767S

MAX767, MAX767T

ELECTRICAL CHARACTERISTICS

(VCC = ON = 5V, GND = PGND = SYNC = 0V, IREF = 0mA, TA= TMIN to TMAX, unless otherwise noted. Typical values are at TA= +25°C.)

Reference Voltage (REF) 3.26 3.30 3.34 V

MAX767T

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 3

100

0.001 0.1 10

EFFICIENCY vs. OUTPUT CURRENT

(1.5A CIRCUIT)

MAX767-01

OUTPUT CURRENT (A)

EFFICIENCY (%)

0.01 1

100

0.001 0.1 10

EFFICIENCY vs. OUTPUT CURRENT

(3A CIRCUIT)

MAX767-02

OUTPUT CURRENT (A)

EFFICIENCY (%)

0.01 1

100

0.001 0.1 10

EFFICIENCY vs. OUTPUT CURRENT

(5A CIRCUIT)

MAX767-03

OUTPUT CURRENT (A)

EFFICIENCY (%)

0.01 1

100

0.001 0.1 10

EFFICIENCY vs. OUTPUT CURRENT

(7A CIRCUIT)

MAX767-04

OUTPUT CURRENT (A)

EFFICIENCY (%)

0.01 1

IDLE-MODE WAVEFORMS

ILOAD = 300mA

3.3V OUTPUT

50mV/div, AC COUPLED

5V/div

5µs/div

100

0.001 0.1 10

EFFICIENCY vs. OUTPUT CURRENT

(10A CIRCUIT)

MAX767-05

OUTPUT CURRENT (A)

EFFICIENCY (%)

0.01 1

1000

0.01

0.001 1 100

SWITCHING FREQUENCY vs.

PERCENT OF FULL LOAD

0.1

MAX767-06

LOAD CURRENT (% FULL LOAD)

SWITCHING FREQUENCY (kHz)

100

0.01 0.1 10

SYNC = REF (300kHz)

PWM-MODE WAVEFORMS

ILOAD = 5A

3.3V OUTPUT

50mV/div, AC COUPLED

5V/div

1µs/div

__________________________________________Typical Operating Characteristics

(Circuit of Figure 1 (5A configuration), VIN = 5V, oscillator frequency = 300kHz, TA= +25°C, unless otherwise noted.)

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

4 _______________________________________________________________________________________

_____________________________Typical Operating Characteristics (continued)

(Circuit of Figure 1 (5A configuration), VIN = 5V, oscillator frequency = 300kHz, TA= +25°C, unless otherwise noted.)

1.5A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

1.5A

3.3V OUTPUT

50mV/div

AC-COUPLED

LOAD CURRENT

3A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

3.3V OUTPUT

50mV/div

AC-COUPLED

LOAD CURRENT

5A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

3.3V OUTPUT

50mV/div

AC-COUPLED

LOAD CURRENT

10A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

10A

3.3V OUTPUT

50mV/div

AC-COUPLED

LOAD CURRENT

7A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

3.3V OUTPUT

50mV/div

AC-COUPLED

LOAD CURRENT

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 5

______________________________________________________________Pin Description

NAME FUNCTION

1CS Current-sense input: +100mV = nominal current-limit level referred to FB.

2SS Soft-start input. Ramp time to full current limit is 1ms/nF of capacitance to GND.

3ON ON/O

—

–control input to disable the PWM. Tie directly to VCC for automatic start-up.

DL Gate-drive output for the low-side synchronous rectifier MOSFET

4–7, 11 GND Low-current analog ground. Feedback reference point for the output.

8REF 3.3V internal reference output. Bypass to GND with 0.22µF minimum capacitor.

9SYNC

Oscillator control/synchronization input. Connect to VCC or GND for 200kHz; connect to REF for

300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition

causes a new cycle to start.

10, 14, 15 VCC Supply voltage input: 4.5V to 5.5V

17 BST Boost capacitor connection (0.1µF)

12 N.C. No internal connection

13 PGND Power ground

18 LX Inductor connection. Can swing 2V below GND without latchup.

19 DH Gate-drive output for the high-side MOSFET

20 FB Feedback and current-sense input for the PWM

OUTPUT

3.3V

GND

MAX767

VCC

REF

BST

PGND

SHUTDOWN

ON/OFF

SMALL-

SIGNAL

SCHOTTKY

0.1µF

0.22µF

(OPTIONAL)

SYNC

10Ω

INPUT

4.5V TO 5.5V

0.01µF

4.7µF

Figure 1. Standard Application Circuit

MAX767

_____Standard Application Circuits

This data sheet shows five predesigned circuits with

output current capabilities from 1.5A to 10A. Many

users will find one of these standard circuits appropri-

ate for their needs. If a standard circuit is used, the

remainder of this data sheet (Detailed Description and

Applications Information and Design Procedure) can

be bypassed.

Figure 1 shows the Standard Application Circuit. Table 1

gives component values and part numbers for five dif-

ferent implementations of this circuit: 1.5A, 3A, 5A, 7A,

and 10A output currents.

Each of these circuits is designed to deliver the full

rated output load current over the temperature range

listed. In addition, each will withstand a short circuit of

several seconds duration from the output to ground. If

the circuit must withstand a continuous short circuit,

refer to the Short-Circuit Duration section for the

required changes.

Layout and Grounding

Good layout is necessary to achieve the designed out-

put power, high efficiency, and low noise. Good layout

includes the use of a ground plane, appropriate com-

ponent placement, and correct routing of traces using

appropriate trace widths. The following points are in

order of decreasing importance.

1. A ground plane is essential for optimum perfor-

mance. In most applications, the circuit will be

located on a multilayer board and full use of the four

or more copper layers is recommended. Use the

top and bottom layers for interconnections and the

inner layers for an uninterrupted ground plane.

2. Because the sense resistance values are similar to

a few centimeters of narrow traces on a printed cir-

cuit board, trace resistance can contribute signifi-

cant errors. To prevent this, Kelvin connect CS and

FB to the sense resistor; i.e., use separate traces

not carrying any of the inductor or load current, as

shown in Figure 2. These signals must be carefully

shielded from DH, DL, BST, and the LX node.

Important: place the sense resistor as close as pos-

sible to and no further than 10mm from the MAX767.

3. Place the LX node components N1, N2, L1, and D2

as close together as possible. This reduces resis-

tive and switching losses and confines noise due to

ground inductance.

4. The input filter capacitor C1 should be less than

10mm away from N1’s drain. The connecting cop-

per trace carries large currents and must be at least

2mm wide, preferably 5mm.

5. Keep the gate connections to the MOSFETs short

for low inductance (less than 20mm long and more

than 0.5mm wide) to ensure clean switching.

6. To achieve good shielding, it is best to keep all

switching signals (MOSFET gate drives DH and DL,

BST, and the LX node) on one side of the board

and all sensitive nodes (CS, FB, and REF) on the

other side.

7. Connect the GND and PGND pins directly to the

ground plane, which should ideally be an inner

layer of a multilayer board.

_______________Detailed Description

Note: The remainder of this document contains the

detailed information necessary to design a circuit that

differs substantially from the five standard application

circuits. If you are using one of the predesigned stan-

dard circuits, the following sections are provided only

for your reading pleasure.

The MAX767 converts a 4.5V to 5.5V input to a 3.3V

output. Its load capability depends on external compo-

nents and can exceed 10A. The 3.3V output is generat-

ed by a current-mode, pulse-width-modulation (PWM)

step-down regulator. The PWM regulator operates at

either 200kHz or 300kHz, with a corresponding trade-

off between somewhat higher efficiency (200kHz) and

smaller external component size (300kHz). The

MAX767 also has a 3.3V, 5mA reference voltage. Fault-

protection circuitry shuts off the output should the refer-

ence lose regulation or the input voltage go below 4V

(nominally).

External components for the MAX767 include two N-

channel MOSFETs, a rectifier, and an LC output filter.

The gate-drive signal for the high-side MOSFET, which

must exceed the input voltage, is provided by a boost

circuit that uses a 0.1µF capacitor. The synchronous

rectifier keeps efficiency high by clamping the voltage

across the rectifier diode. An external low-value cur-

rent-sense resistor sets the maximum current limit, pre-

venting excessive inductor current during start-up or

under short-circuit conditions. An optional external

capacitor sets the programmable soft-start, reducing

in-rush surge currents upon start-up and providing

adjustable power-up time.

The PWM regulator is a direct-summing type, lacking a

traditional integrator-type error amplifier and the phase

shift associated with it. It therefore does not require

external feedback-compensation components, as long

as you follow the ESR guidelines in the Applications

Information and Design Procedure sections.

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

6 _______________________________________________________________________________________

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 7

Part 1.5A Circuit 3A Circuit 5A Circuit 7A Circuit 10A Circuit

L1 10µH

Sumida CDR74B-100

5µH

Sumida CDR125

DRG# 4722-JPS-001

3.3µH

Coilcraft

DO3316-332

2.1µH, 5mΩ

Coiltronics

CTX03-12338-1

1.5µH, 3.5mΩ

Coiltronics

CTX03-12357-1

0.04Ω

IRC LR2010-01-R040

or DD WSL-2512-R040

0.02Ω

IRC LR2010-01-R020

or DD WSL-2512-R020

0.012Ω

DD WSL-2512-R012

or 2 x 0.025Ω

IRC LR2010-01-R025

(in parallel)

3 x 0.025Ω

IRC LR2010-01-R025

or DD WSL-2512-R025

(in parallel)

3 x 0.020Ω

IRC LR2010-01-R020

or 2 x 0.012Ω

DD WSL-2512-R012

(in parallel)

N1,

International

Rectifier IRF7101,

Siliconix Si9936DY

or Motorola

MMDF3N03HD

(dual N-channel)

Siliconix Si9410DY,

International

Rectifier IRF7101

or Motorola

MMDF3N03HD

(both FETs in parallel)

Motorola

MTD20N03HDL

Motorola

MTD75N03HDL

(N1)

MTD20N03HDL

(N2)

Motorola

MTD75N03HDL

C1 47µF, 20V

AVX TPSD476K020R

2 x 47µF, 20V

AVX TPSD476K020R

220µF, 10V

Sanyo

OS-CON 10SA220M

2 x 100µF, 10V

Sanyo

OS-CON 10SA100M

2 x 220µF, 10V

Sanyo

OS-CON 10SA220M

220µF, 6.3V

Sprague

595D227X06R3D2B

2 x 150µF, 10V

Sprague

595D157X0010D7T

2 x 220µF, 10V

Sanyo

OS-CON 10SA220M

2 x 220µF, 10V

Sanyo

OS-CON 10SA220M

4 x 220µF, 10V

Sanyo

OS-CON 10SA220M

1N5817

Nihon EC10QS02,

or Motorola

MBRS120T3

1N5817

Nihon EC10QS02,

or Motorola

MBRS120T3

1N5820

Nihon NSQ03A02,

or Motorola

MBRS340T3

1N5820

Nihon NSQ03A02,

or Motorola

MBRS340T3

1N5820

Nihon NSQ03A02,

or Motorola

MBRS340T3

Temp.

Range to +85°C to +85°C to +85°C to +85°C to +85°C

Table 1. Component Values

Table 2. Component Suppliers

Company Factory Fax [Country Code] USA Telephone

AVX [1] (803) 626-3123 (803) 946-0690

(800) 282-4975

Coilcraft [1] (847) 639-1469 (847) 639-6400

Coiltronics [1] (561) 241-9339 (561) 241-7876

DD [1] (402) 563-6418 (402) 564-3131

IRC [1] (512) 992-3377 (512) 992-7900

International

Rectifier [1] (310) 322-3332 (310) 322-3331

Nihon [81] 3-3494-7414 (805) 867-2555

Sanyo [81] 7-2070-1174 (619) 661-6835

Siliconix [1] (408) 970-3950 (408) 988-8000

Sprague [1] (603) 224-1430 (603) 224-1961

Sumida [81] 3-3607-5144 (847) 956-0666

Motorola [1] (602) 994-6430 (602) 303-5454

MAX767

The main gain block is an open-loop comparator that

sums four signals: output voltage error signal, current-

sense signal, slope-compensation ramp, and the 3.3V

reference. This direct-summing method approaches

the ideal of cycle-by-cycle control of the output voltage.

Under heavy loads, the controller operates in full PWM

mode. Every pulse from the oscillator sets the output

latch and turns on the high-side switch for a period

determined by the duty factor (approximately

VOUT / VIN).

As the high-side switch turns off, the synchronous recti-

fier latch is set; 60ns later, the low-side switch turns on.

The low-side switch stays on until the beginning of the

next clock cycle (in continuous-conduction mode) or

until the inductor current reaches zero (in discontinu-

ous-conduction mode). Under fault conditions where

the inductor current exceeds the 100mV current-limit

threshold, the high-side latch resets and the high-side

switch turns off.

At light loads, the inductor current fails to exceed the

25mV threshold set by the minimum-current compara-

tor. When this occurs, the PWM goes into Idle-Mode™,

skipping most of the oscillator pulses to reduce the

switching frequency and cut back switching losses.

The oscillator is effectively gated off at light loads

because the minimum-current comparator immediately

resets the high-side latch at the beginning of each

cycle, unless the FB signal falls below the reference

voltage level.

Soft-Start

Connecting a capacitor from the soft-start pin (SS) to

ground allows a gradual build-up of the 3.3V output

after power is applied or ON is driven high. When ON is

low, the soft-start capacitor is discharged to GND.

When ON is driven high, a 4µA constant current source

charges the capacitor up to 4V. The resulting ramp volt-

age on SS linearly increases the current-limit compara-

tor set-point, increasing the duty cycle to the external

power MOSFETs. With no soft-start capacitor, the full

output current is available within 10µs (see Applications

Information and Design Procedure section).

Synchronous Rectifier

Synchronous rectification allows for high efficiency by

reducing the losses associated with the Schottky rectifi-

er. Also, the synchronous-rectifier MOSFET is neces-

sary for correct operation of the MAX767’s boost gate-

drive supply.

When the external power MOSFET (N1) turns off, ener-

gy stored in the inductor causes its terminal voltage to

reverse instantly. Current flows in the loop formed by

the inductor (L1), Schottky diode (D2), and the load—

an action that charges up the output filter capacitor

(C2). The Schottky diode has a forward voltage of

about 0.5V which, although small, represents a signifi-

cant power loss and degrades efficiency. The synchro-

nous-rectifier MOSFET parallels the diode and is turned

on by DL shortly after the diode conducts. Since the

synchronous rectifier’s on resistance (rDS(ON)) is very

low, the losses are reduced. The synchronous-rectifier

MOSFET is turned off when the inductor current falls to

zero.

The MAX767’s internal break-before-make timing

ensures that shoot-through (both external switches

turned on at the same time) does not occur. The

Schottky rectifier conducts during the time that neither

MOSFET is on, which improves efficiency by preventing

the synchronous-rectifier MOSFET’s lossy body diode

from conducting.

The synchronous rectifier works under all operating

conditions, including discontinuous-conduction mode

and idle-mode.

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

8 _______________________________________________________________________________________

MAX767

SENSE RESISTOR

MAIN CURRENT PATH

FAT, HIGH-CURRENT TRACES

Figure 2. Kelvin Connections for the Current-Sense Resistor

Idle-Mode is a trademark of Maxim Integrated Products.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 9

SHOOT-

THROUGH

CONTROL

60kHz

LPF

MINIMUM

CURRENT

(IDLE-MODE)

25mV

QLEVEL

SHIFT

MAIN PWM

COMPARATOR

LEVEL

SHIFT

CURRENT

LIMIT

VCC

30R

3.3V

4µA

SYNCHRONOUS

RECTIFIER CONTROL

0mV TO 100mV

BST

VCC

PGND

SLOPE COMP

200kHz/300kHz

OSCILLATOR

+3.3V

REFERENCE

2.8V

REF

VCC

FAULT

SYNC

MAX767

Figure 3. MAX767 Block Diagram

MAX767

Gate-Driver Boost Supply

Gate-drive voltage for the high-side N-channel switch is

generated with the flying-capacitor boost circuit shown

in Figure 4. The capacitor (C3) is alternately charged

from the 5V input via the diode (D1) and placed in par-

allel with the high-side MOSFET’s gate-source termi-

nals. On start-up, the synchronous rectifier (low-side)

MOSFET (N2) forces LX to 0V and charges the BST

capacitor to 5V. On the second half-cycle, the PWM

turns on the high-side MOSFET (N1); it does this by

closing an internal switch between BST and DH, which

connects the capacitor to the MOSFET gate. This pro-

vides the necessary enhancement voltage to turn on

the high-side switch, an action that “boosts” the 5V

gate-drive signal above the input voltage.

Ringing seen at the high-side MOSFET gates (DH) in

discontinuous-conduction mode (light loads) is a natur-

al operating condition. It is caused by the residual

energy in the tank circuit, formed by the inductor and

stray capacitance at the LX node. The gate-driver neg-

ative rail is referred to LX, so any ringing there is direct-

ly coupled to the gate-drive supply.

Modes of Operation

PWM Mode

Under heavy loads—over approximately 25% of full

load—the supply operates as a continuous-current

PWM supply (see Typical Operating Characteristics).

The duty cycle, %ON, is approximately:

VOUT

%ON = ________

VIN

Current flows continuously in the inductor: first, it ramps

up when the power MOSFET conducts; second, it

ramps down during the flyback portion of each cycle as

energy is put into the inductor and then discharged into

the load. Note that the current flowing into the inductor

when it is being charged is also flowing into the load,

so the load is continuously receiving current from the

inductor. This minimizes output ripple and maximizes

inductor use, allowing very small physical and electrical

sizes. Output ripple is primarily a function of the filter

capacitor’s effective series resistance (ESR), and is

typically under 50mV (see Design Procedure section).

Idle-Mode

Under light loads (<25% of full load), the MAX767

enhances efficiency by turning the drive voltage on and

off for only a single clock period, skipping most of the

clock pulses entirely. Asynchronous switching, seen as

“ghosting” on an oscilloscope, is thus a normal operat-

ing condition whenever the load current is less than

approximately 25% of full load.

At certain input voltage and load conditions, a transition

region exists where the controller can pass back and

forth from idle-mode to PWM mode. In this situation,

short pulse bursts occur, which make the current wave-

form look erratic but do not materially affect the output

ripple. Efficiency remains high.

Current Limiting

The voltage between CS and FB is continuously moni-

tored. An external, low-value shunt resistor is connect-

ed between these pins, in series with the inductor,

allowing the inductor current to be continuously mea-

sured throughout the switching cycle. Whenever this

voltage exceeds 100mV, the drive voltage to the exter-

nal high-side MOSFET is cut off. This protects the MOS-

FET, the load, and the input supply in case of short cir-

cuits or temporary load surges. The current-limiting

resistance is typically 20mΩfor 3A.

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

10 ______________________________________________________________________________________

LEVEL

TRANSLATOR

PWM

VIN

BST

VCC

VCC MAX767

Figure 4. Boost Supply for High-Side Gate Driver

Oscillator Frequency

The SYNC input controls the oscillator frequency.

Connecting SYNC to GND or to VCC selects 200kHz

operation; connecting it to REF selects 300kHz opera-

tion. SYNC can also be driven with an external 240kHz

to 350kHz CMOS/TTL source to synchronize the inter-

nal oscillator. Normally, 300kHz operation is chosen to

minimize the inductor and output filter capacitor sizes,

but 200kHz operation may be chosen for a small (about

1%) increase in efficiency at heavy loads.

Internal Reference

The internal 3.3V bandgap reference (REF) remains

active, even when the switching regulator is turned off.

It can furnish up to 5mA, and can be used to supply

memory keep-alive power or for other purposes.

Bypass REF to GND with 0.22µF, plus 1µF/mA of load

current.

Applications Information and

__________________Design Procedure

Most users will be able to work with one of the standard

application circuits; others may want to implement a

circuit with an output current rating that lies between or

beyond the standard values.

If you want an output current level that lies between two

of the standard application circuits, you can interpolate

many of the component values from the values given

for the two circuits. These components include the

input and output filter capacitors, the inductor, and the

sense resistor. The capacitors must meet ESR and rip-

ple current requirements (see Input Filter Capacitor and

Output Filter Capacitor sections). The inductor must

meet the required current rating (see Inductor section).

You may use the rectifier and MOSFETs specified for

the circuit with the greater output current capability, or

choose a new rectifier and MOSFETs according to the

requirements detailed in the Rectifier and MOSFET

Switches sections. For more complete information, or

for output currents in excess of 10A, refer to the design

information in the following sections.

Inductor, L1

Three inductor parameters are required: the inductance

value (L), the peak inductor current (ILPEAK), and the

coil resistance (RL). The inductance is:

1.32

L1 = ______________

f x IOUT x LIR

where:

f = switching frequency, normally 300kHz

IOUT = maximum 3.3V DC load current (A)

LIR = ratio of inductor peak-to-peak AC

current to average DC load current,

typically 0.3.

A higher LIR value allows smaller inductance, but

results in higher losses and ripple.

The highest peak inductor current (ILPEAK) equals the

DC load current (IOUT) plus half the peak-to-peak AC

inductor current (ILPP). The peak-to-peak AC inductor

current is typically chosen as 30% of the maximum DC

load current, so the peak inductor current is 1.15 x IOUT.

The peak inductor current at any load is given by:

1.32

ILPEAK = IOUT + __________

2 x f x L1

The coil resistance should be as low as possible,

preferably in the low milliohms. The coil is effectively in

series with the load at all times, so the wire losses alone

are approximately:

Power Loss = IOUT2x RL

In general, select a standard inductor that meets the L,

ILPEAK, and RLrequirements. If a standard inductor is

unavailable, choose a core with an LI2parameter

greater than L x ILPEAK2, and use the largest wire that

will fit the core.

Current-Sense Resistor, R1

The current-sense resistor must carry the peak current

in the inductor, which exceeds the full DC load current.

The internal current limiting starts when the voltage

across the sense resistors exceeds 100mV nominally,

80mV minimum. Use the minimum value to ensure ade-

quate output current capability: R1 = 80mV / ILPEAK.

The low VIN/VOUT ratio creates a potential problem with

start-up under full load or with load transients from no-

load to full load. If the supply is subjected to these con-

ditions, reduce the sense resistor:

70mV

R1 = ———

ILPEAK

Since the sense-resistance values are similar to a few

centimeters of narrow traces on a printed circuit board,

trace resistance can contribute significant errors. To

prevent this, Kelvin connect the CS and FB pins to the

sense resistors; i.e., use separate traces not carrying

any of the inductor or load current, as shown in Figure 2.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 11

MAX767

Place R1 as close as possible to the MAX767, prefer-

ably less than 10mm. Run the traces at minimum spac-

ing from one another. If they are longer than 20mm,

bypass CS to FB with a 1nF capacitor placed as close

as possible to these pins. The wiring layout for these

traces is critical for stable, low-ripple outputs (see

Layout and Grounding section).

Input Filter Capacitor, C1

Use at least 6µF per watt of output power for C1. If the

5V input is some distance away or comes through a PC

bus, greater capacitance may be desirable to improve

the load-transient response. Use a low-ESR capacitor

located no further than 10mm from the MOSFET switch

(N1) to prevent ringing. The ripple current rating must

be at least IRMS = 0.5 x IOUT. For high-current applica-

tions, two or more capacitors in parallel may be needed

to meet these requirements.

The ESR of C1 is effectively in series with the input. The

resistive dissipation of C1, IRMS2x ESRC1, can signifi-

cantly impact the circuit’s efficiency.

Output Filter Capacitor, C2

The output filter capacitor determines the loop stability,

output voltage ripple, and output load-transient

response.

Stability

To ensure stability, stay above the minimum capaci-

tance value and below the maximum ESR value. These

values are:

3Ω

C2 > —— µF

and

ESRC2 < R1

Be sure to satisfy both these requirements. To achieve

the low ESR required, it may be appropriate to parallel

two or more capacitors and/or use a total capacitance

2 or 3 times larger than the calculated minimum.

Output Ripple

The output ripple in continuous-conduction mode is:

VOUT(RPL) = IOUT(max) x LIR x

(ESRC2 + ———————)

2 x πx f x C2

where f is the switching frequency (200kHz or 300kHz).

In idle-mode, the ripple has a capacitive and a resistive

component:

0.0004 x L

VOUT(RPL) (C) = _____________ x 0.89 Volts

R12x C2

0.02 x ESRC2

VOUT(RPL) (R) = _____________

The total ripple, VOUT(RPL), can be approximated as

follows:

VOUT(RPL) (R) < 0.5 VOUT(RPL) (C)

then

VOUT(RPL) = VOUT(RPL)(C)

otherwise

VOUT(RPL) = 0.5 VOUT(RPL) (C) +

VOUT(RPL) (R)

Load-Transient Performance

In response to a large step increase in load current, the

output voltage will sag for several microseconds unless

C2 is increased beyond the values that satisfy the

above requirements. Note that an increase in capaci-

tance is all that’s required to improve the transient

response, and that the ESR requirements don’t change.

Therefore, the added capacitance can be supplied by

an additional low-cost bulk capacitor in parallel with the

normal low-ESR switching-regulator capacitor. The

equation for voltage sag under a step load change is:

ISTEP2x L

VSAG = ________________________________

2 x C2 x (VIN(min) x DMAX - 3.3V)

where DMAX is the maximum duty cycle. Higher duty

cycles are possible when the oscillator frequency is

reduced to 200kHz, since fixed propagation delays

through the PWM comparator become a lesser part of

the whole period. The tested worst-case limit for DMAX

is 92% at 200kHz or 89% at 300kHz. Lower inductance

values can reduce the filter capacitance requirement,

but only at the expense of increased output ripple (due

to higher peak currents).

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

12 ______________________________________________________________________________________

RC Filter for VCC

R2 and C4 form a lowpass filter to remove switching

noise from the VCC input to the MAX767. C4 must have

fairly low ESR (<5Ω). Switching noise can interfere with

proper output voltage regulation, resulting in an exces-

sive output voltage decrease (>100mV) at full load.

Overheating during soldering can damage the surface-

mount capacitors specified for C4, causing the regula-

tion problems described above. Take care to heat the

capacitor for as short a time as possible, especially if it

is soldered by hand.

Rectifier, D2

Use a 1N5817 or similar Schottky diode for applications

up to 3A, or a 1N5820 for up to 10A. Surface-mount

equivalents are available from N.I.E.C. with part num-

bers EC10QS02 and NSQ03A02, or from Motorola with

part numbers MBRS120T3 and MBRS320T3. D2 must

be a Schottky diode to prevent the lossy MOSFET body

diode from turning on.

Soft-Start

A capacitor connected from GND to SS causes the

supply’s current-limit level to ramp up slowly. The ramp

time to full current limit is approximately 1ms for every

nF of capacitance on SS, with a minimum value of

10µs. Typical values for the soft-start capacitor are in

the 10nF to 100nF range; a 5V rating is sufficient.

The time required for the output voltage to ramp up to

its rated value depends upon the output load, and is

not necessarily the same as the time it takes for the cur-

rent limit to reach full capacity.

Duty Cycle

The duty cycle for the high-side MOSFET (N1) in con-

tinuous-conduction mode is:

100% x ( VOUT + VN2)

___________________

VIN - VN1

where:

VOUT = 3.3V

VIN = 5V

VN1 and VN2 = ILOAD x rDS(ON) for each MOSFET.

It is apparent that, in continuous-conduction mode, N1

will conduct for about twice the time as N2. Under short-

circuit conditions, however, N2 can conduct as much

90% of the time. If there is a significant chance of short

circuiting the output, select N2 to handle the resulting

duty cycle (see Short-Circuit Duration section).

MOSFET Switches, N1 and N2

The two N-channel MOSFETs must be “logic-level”

FETs; that is, they must be fully on (have low

rDS(ON)) with only 4V gate-source drive voltage. For

high-current applications, FETs with low gate-

threshold voltage specifications (i.e., maximum

VGS(TH) = 2V rather than 3V) are preferred. In addi-

tion, they should have low total gate charge (<70nC)

to minimize switching losses.

For output currents in excess of the five standard appli-

cation circuits, placing MOSFETs with very low gate

charge in parallel increases output current and lowers

resistive losses. N2 does not normally require the same

current capacity as N1 because it conducts only about

33% of the time, while N1 conducts about 66% of the

time.

Short-Circuit Duration

At their highest rated temperatures (+70°C or +85°C),

each of the five standard application circuits will with-

stand a short circuit of several seconds duration. In

most cases, the MAX767 will be used in applications

where long-term short circuiting of the output is unlikely.

If it is desirable for the circuit to withstand a continuous

short circuit, the MOSFETs must be able to dissipate

the required power. This depends on physical factors

such as the mounting of the transistor, any heat-

sinking used, and ventilation provided, as well as the

actual current the transistor must deliver. The short-

circuit current is approximately 100mV / R1, but may

vary by ±20%.

Cautious design requires that the transistors

withstand the maximum possible current, which is

ISC = 120mV / R1. N1 and N2 must withstand this

current scaled by their maximum duty factors. The

maximum duty factor for N1 occurs under high-

load (but not short-circuit) conditions, and is approxi-

mately VOUT / VIN(min) or about 0.7. The max-

imum duty factor for N2 occurs during short-circuit

conditions and is:

ISC x rDS(ON)N2

1 - —————————————

VIN(max) - ISC x rDS(ON)N1

which can exceed 0.9. The total power dissipated in

both MOSFETs together is ISC2x rDS(ON).

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 13

MAX767

Proper circuit operation requires that the short-circuit

current be at least ILOAD x (1 + LIR / 2). However, the

standard application circuits are designed for a short-

circuit current slightly in excess of this amount. This

excess design current guarantees proper start-up

under constant full-load conditions and proper full-load

transient response, and is particularly necessary with

low input voltages. If the circuit will not be subjected to

full-load transients or to loads approaching the full-load

at start-up, you can decrease the short-circuit current

by increasing R1, as described in the Current-Sense

Resistor section. This may allow use of MOSFETs with a

lower current-handling capability.

Heavy-Load Efficiency

Losses due to parasitic resistances in the switches,

coil, and sense resistor dominate at high load-current

levels. Under heavy loads, the MAX767 operates deep

in the continuous-conduction mode, where there is a

large DC offset to the inductor current (plus a small

sawtooth AC component) (see Inductor section). This

DC current is exactly equal to the load current, a fact

which makes it easy to estimate resistive losses via the

simplifying assumption that the total inductor current is

equal to this DC offset current. The major loss mecha-

nisms under heavy loads, in usual order of importance,

are:

•I

2R losses

• gate-charge losses

• diode-conduction losses

• transition losses

• capacitor-ESR losses

• losses due to the operating supply current of the IC.

Inductor-core losses, which are fairly low at heavy

loads because the AC component of the inductor cur-

rent is small, are not accounted for in this analysis.

POUT

Efficiency = ______ x 100% =

POUT

_______________ x 100%

POUT + PDTOTAL

PDTOTAL = PD(I2R) + PDGATE + PDDIODE +

PDTRAN + PDCAP + PDIC

I2R Losses

PD(I2R) = resistive loss = (ILOAD2) x

(RCOIL + rDS(ON) + R1)

where RCOIL is the DC resistance of the coil and

rDS(ON) is the drain-source on resistance of the MOS-

FET. Note that the rDS(ON) term assumes that identical

MOSFETs are employed for both the synchronous recti-

fier and high-side switch, because they time-share the

inductor current. If the MOSFETs are not identical, esti-

mate losses by averaging the two individual rDS(ON)

terms according to their duty factors: 0.66 for N1 and

0.34 for N2.

Gate-Charge Losses

PDGATE = gate driver loss = qGx f x 5V

where qGis the sum of the gate charge for low- and

high-side switches. Note that gate-charge losses are

dissipated in the IC, not the MOSFETs, and therefore

contribute to package temperature rise. For a pair of

matched MOSFETs, qGis simply twice the gate capaci-

tance of a single MOSFET (a data sheet specification).

Diode Conduction Losses

PDDIODE = diode conduction losses =

ILOAD x VDx tDx f

where VDis the forward voltage of the Schottky diode

at the output current, tDis the diode’s conduction time

(typically 110ns), and f is the switching frequency.

Transition Losses

PDTRAN = transition loss =

VIN2x CRSS x ILOAD x f

______________________

IDRIVE

where CRSS is the reverse transfer capacitance of the

high-side MOSFET (a data sheet parameter), f is the

switching frequency, and IDRIVE is the peak current

available from the high-side gate driver output (approx-

imately 1A).

Additional switching losses are introduced by other

sources of stray capacitance at the switching node,

including the catch-diode capacitance, coil interwind-

ing capacitance, and low-side switch drain capaci-

tance, and are given as PDSW = VIN2x CSTRAY x f, but

these are usually negligible compared to CRSS losses.

The low-side switch introduces only tiny switching loss-

es, since its drain-source voltage is already low when it

turns on.

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

14 ______________________________________________________________________________________

Capacitor ESR Losses

PDCAP = capacitor ESR loss = IRMS2x ESR

where IRMS = RMS AC input current, approximately

ILOAD / 2.

Note that losses in the output filter capacitors are small

when the circuit is heavily loaded, because the current

into the capacitor is not chopped. The output capacitor

sees only the small AC sawtooth ripple current. Ensure

that the input bypass capacitor has a ripple current rat-

ing that exceeds the value of IRMS.

IC Supply-Current Losses

PDIC is the quiescent power dissipation of the IC and is

5V times the quiescent supply current (a data sheet

parameter), or about 5mW.

Light-Load Efficiency

Under light loads, the PWM will operate in discontinu-

ous-conduction mode, where the inductor current dis-

charges to zero at some point during each switching

cycle. New loss mechanisms, insignificant at heavy

loads, begin to become important. The basic difference

is that in discontinuous mode, the AC component of the

inductor current is large compared to the load current.

This increases losses in the core and in the output filter

capacitors. Ferrite cores are recommended over pow-

dered-material types for best light-load efficiency.

At light loads, the inductor delivers triangular current

pulses rather than the nearly square waves found in

continuous-conduction mode. These pulses ramp up to

a point set by the idle-mode current comparator, which

is internally fixed at approximately 25% of the full-scale

current-limit level. This 25% threshold provides an opti-

mum balance between low-current efficiency and out-

put voltage noise (the efficiency curve would actually

look better with this threshold set at about 45%, but the

output noise would be too high).

____Additional Application Circuits

High-Accuracy Power Supplies

The standard application circuit’s accuracy is dominat-

ed by reference voltage error (±1.8%) and load regula-

tion error (-2.5%). Both of these parameters can be

improved as shown in Figures 5 and 6. Both circuits

rely on an external integrator amplifier to increase the

DC loop gain in order to reduce the load regulation

error to 0.1%. Reference error is improved in the first

circuit by employing a version of the MAX767 (“T”

grade) which has a ±1.2% reference voltage tolerance.

Reference error of the second circuit is further

improved by substituting a highly accurate external ref-

erence chip (MAX872), which contributes ±0.38% total

error over temperature.

These two circuits were designed with the latest gener-

ation of dynamic-clock µPs in mind, which place great

demands on the transient-response performance of the

power supply. As the µP clock starts and stops, the

load current can change by several amps in less than

100ns. This tremendous ∆i/∆t can cause output voltage

overshoot or sag that results in the CPU VCC going out

of tolerance unless the power supply is carefully

designed and located close to the CPU. These circuits

have excellent dynamic response and low ripple, with

transient excursions of less than 40mV under zero to

full-load step change. In particular, these two circuits

support the “VR” (voltage regulator) version of the Intel

P54C Pentium™ CPU, which requires that its supply

voltage, including noise and transient errors, be within

the 3.30V to 3.45V range.

To configure these circuits for a given load current

requirement, substitute standard components from

Table 1 for the power switching elements (N1, N2, L1,

C1, C2) or use the Design Procedure. R1 can also be

taken from Table 1, but must be adjusted approximate-

ly 10% higher in order to maintain the correct current-

limit threshold. This increased value is due to the 0.9

gain factor introduced by the H-bridge resistor divider

(R3–R6).

If the remote sense line must sense the output voltage

on the far side of a connector or jumper that has the

possibility of becoming disconnected while the power

supply is operating, an additional 10kΩresistor should

connect the sense line to the output voltage in the con-

nector’s power-supply side in order to prevent acciden-

tal overvoltage at the CPU.

For applications that are powered from a fixed +12V or

battery input rather than from +5V, use a MAX797 IC

instead of the MAX767. The MAX797 is capable of

accepting inputs up to 30V. See the MAX796–MAX799

data sheet for a high-accuracy circuit schematic.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 15

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

16 ______________________________________________________________________________________

3.38V OUTPUT

3.427V MAX

3.330V MIN

GND

MAX767T

VCC

REF

BST

PGND

SHUTDOWN

ON/OFF

C2R1

0.01µF

SYNC

10Ω

4.7µF

TO MAX767

VCC

332k, 1%

10k

R10

8.06k, 1%

1k,

10k,

10µF CERAMIC

(LOCATE AT

µP PINS)

R11

5.1k

MIN

LOAD

0.01µF

0.1µF

0.22µF

C10

0.01µF

(OPTIONAL)

620pF

INPUT

4.75V TO 5.5V

VOUT = VREF ( R10 + 1)

MAX495

REMOTE SENSE LINE

330k

Figure 5. High-Accuracy CPU Power Supply with Internal Reference

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 17

3.38V OUTPUT

3.408V MAX

3.369V MIN

GND

MAX767

VCC

REF

BST

PGND

SHUTDOWN

ON/OFF

C2R1

0.01µF

SYNC

20Ω

22µF

332k, 0.1%

10k

R10

118k, 0.1%

1k,

10k,

10µF CERAMIC

(LOCATE AT

µP PINS)

R11

5.1k

MIN

LOAD

0.01µF

0.1µF

C10

0.01µF

(OPTIONAL)

1000pF

INPUT

4.75V TO 5.5V

VOUT = VREF ( R10 + 1)

MAX495

REMOTE SENSE LINE

330k

GND

VIN

VOUT

MAX872

TO MAX767

VCCC9

0.22µF

Figure 6. High-Accuracy CPU Power Supply with External Reference

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

18 ______________________________________________________________________________________

___________________Chip TopographyOrdering Information (continued)

TRANSISTOR COUNT: 1294

SUBSTRATE CONNECTED TO GND

BST

GND

REF

GND

VCC

0.181"

(4.597mm)

0.109"

(2.769mm)

GND

SYNC

PGND

VCC

SS CS FB DH

3.3±1.2%20 SSOP-40°C to +85°CMAX767TEAP

3.6

3.45

3.3

VOUT

(V)

±1.8%

REF

TOL

20 SSOP-40°C to +85°CMAX767SEAP

20 SSOP-40°C to +85°CMAX767REAP

20 SSOP-40°C to +85°CMAX767EAP

PIN-

PACKAGE

TEMP. RANGEPART

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 19

________________________________________________________Package Information

DIM

MIN

0.068

0.002

0.010

0.005

0.278

0.205

0.301

0.022

0˚

MAX

0.078

0.008

0.015

0.009

0.289

0.212

0.311

0.037

8˚

MIN

1.73

0.05

0.25

0.13

7.07

5.20

7.65

0.55

0˚

MAX

1.99

0.21

0.38

0.22

7.33

5.38

7.90

0.95

8˚

INCHES MILLIMETERS

20-PIN PLASTIC

SHRINK

SMALL-OUTLINE

PACKAGE

A1 C

0.127mm

0.004in.

0.65 BSC0.0256 BSC

21-0003A