LM5574MTX - National Semiconductor

LM5574

LM5574/LM5574Q SIMPLE SWITCHER® 75V, 0.5A Step-Down Switching Regulator

Literature Number: SNVS478E

LM5574/LM5574Q

March 23, 2009

SIMPLE SWITCHER® 75V, 0.5A Step-Down Switching

Regulator

General Description

The LM5574 is an easy to use SIMPLE SWITCHER® buck

regulator which allows design engineers to design and opti-

mize a robust power supply using a minimum set of compo-

nents. Operating with an input voltage range of 6 - 75V, the

LM5574 delivers 0.5A of continuous output current with an

integrated 750mΩ N-Channel MOSFET. The regulator uti-

lizes an Emulated Current Mode architecture which provides

inherent line regulation, tight load transient response, and

ease of loop compensation without the usual limitation of low-

duty cycles associated with current mode regulators. The

operating frequency is adjustable from 50kHz to 500kHz to

allow optimization of size and efficiency. To reduce EMI, a

frequency synchronization pin allows multiple IC’s from the

LM(2)557x family to self-synchronize or to synchronize to an

external clock. The LM5574 guarantees robustness with cy-

cle-by-cycle current limit, short-circuit protection, thermal

shut-down, and remote shut-down. The device is available in

a TSSOP-16 package. The LM5574 is supported by the full

suite of WEBENCH® On-Line design tools.

Features

■LM5574Q is an Automotive Grade product that is AEC-

Q100 grade 1 qualified (−40°C to + 125°C operating

junction temperature)

■Integrated 75V, 750mΩ N-channel MOSFET

■Ultra-wide input voltage range from 6V to 75V

■Adjustable output voltage as low as 1.225V

■1.5% feedback reference accuracy

■Operating frequency adjustable between 50kHz and

500kHz with single resistor

■Master or slave frequency synchronization

■Adjustable soft-start

■Emulated current mode control architecture

■Wide bandwidth error amplifier

■Built-in protection

■Automotive Grade product datasheet that is AEC-Q100

grade 0 qualified is available upon request

(−40°C to + 150°C operating junction temperature)

Package

■TSSOP-16

Applications

■Automotive

■Industrial

Simplified Application Schematic

20212001

WEBENCH® is a registered trademark of National Semiconductor Corporation.

LM5574/LM5574Q SIMPLE SWITCHER® 75V, 0.5A Step-Down Switching Regulator

Connection Diagram

20212002

Top View

16-Lead TSSOP

Ordering Information

Order Number Package Type NSC Package Drawing Supplied As Features

LM5574MT TSSOP-16 MTC16 92 Units in Rail

LM5574MTX TSSOP-16 MTC16 2500 Units on Tape and

Reel

LM5574QMT TSSOP-16 MTC16 92 Units in Rail AEC-Q100 Grade 1

qualified. Automotive

Grade Production Flow *

LM5574QMTX TSSOP-16 MTC16 2500 Units on Tape and

Reel

* Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.

Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified

with the letter Q. For more information go to http://www.national.com/automotive.

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LM5574/LM5574Q

Pin Descriptions

Pin(s) Name Description Application Information

1 VCC Output of the bias regulator Vcc tracks Vin up to 9V. Beyond 9V, Vcc is regulated to 7

Volts. A 0.1uF to 1uF ceramic decoupling capacitor is

required. An external voltage (7.5V – 14V) can be applied

to this pin to reduce internal power dissipation.

2 SD Shutdown or UVLO input If the SD pin voltage is below 0.7V the regulator will be in a

low power state. If the SD pin voltage is between 0.7V and

1.225V the regulator will be in standby mode. If the SD pin

voltage is above 1.225V the regulator will be operational. An

external voltage divider can be used to set a line

undervoltage shutdown threshold. If the SD pin is left open

circuit, a 5µA pull-up current source configures the regulator

fully operational.

3 Vin Input supply voltage Nominal operating range: 6V to 75V

4 SYNC Oscillator synchronization input or output The internal oscillator can be synchronized to an external

clock with an external pull-down device. Multiple LM5574

devices can be synchronized together by connection of their

SYNC pins.

5 COMP Output of the internal error amplifier The loop compensation network should be connected

between this pin and the FB pin.

6 FB Feedback signal from the regulated

output

This pin is connected to the inverting input of the internal

error amplifier. The regulation threshold is 1.225V.

7 RT Internal oscillator frequency set input The internal oscillator is set with a single resistor, connected

between this pin and the AGND pin.

8 RAMP Ramp control signal An external capacitor connected between this pin and the

AGND pin sets the ramp slope used for current mode

control. Recommended capacitor range 50pF to 2000pF.

9 AGND Analog ground Internal reference for the regulator control functions

10 SS Soft-start An external capacitor and an internal 10µA current source

set the time constant for the rise of the error amp reference.

The SS pin is held low during standby, Vcc UVLO and

thermal shutdown.

11 OUT Output voltage connection Connect directly to the regulated output voltage.

12 PGND Power ground Low side reference for the PRE switch and the IS sense

resistor.

13 IS Current sense Current measurement connection for the re-circulating

diode. An internal sense resistor and a sample/hold circuit

sense the diode current near the conclusion of the off-time.

This current measurement provides the DC level of the

emulated current ramp.

14 SW Switching node The source terminal of the internal buck switch. The SW pin

should be connected to the external Schottky diode and to

the buck inductor.

15 PRE Pre-charge assist for the bootstrap

capacitor

This open drain output can be connected to SW pin to aid

charging the bootstrap capacitor during very light load

conditions or in applications where the output may be pre-

charged before the LM5574 is enabled. An internal pre-

charge MOSFET is turned on for 250ns each cycle just prior

to the on-time interval of the buck switch.

16 BST Boost input for bootstrap capacitor An external capacitor is required between the BST and the

SW pins. A 0.022µF ceramic capacitor is recommended.

The capacitor is charged from Vcc via an internal diode

during the off-time of the buck switch.

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LM5574/LM5574Q

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN to GND 76V

BST to GND 90V

PRE to GND 76V

SW to GND (Steady State) -1.5V

BST to VCC 76V

SD, VCC to GND 14V

BST to SW 14V

OUT to GND Limited to Vin

SYNC, SS, FB, RAMP to GND 7V

ESD Rating (Note 2)

Human Body Model 2kV

Storage Temperature Range -65°C to +150°C

Operating Ratings (Note 1)

VIN 6V to 75V

Operation Junction Temperature −40°C to + 125°C

Electrical Characteristics Specifications with standard typeface are for TJ = 25°C, and those with boldface type

apply over full Operating Junction Temperature range. VIN = 48V, RT = 32.4kΩ unless otherwise stated. (Note 3)

Symbol Parameter Conditions Min Typ Max Units

STARTUP REGULATOR

VccReg Vcc Regulator Output 6.85 7.15 7.45 V

Vcc LDO Mode turn-off 9 V

Vcc Current Limit Vcc = 0V 25 mA

VCC SUPPLY

Vcc UVLO Threshold (Vcc increasing) 5.03 5.35 5.67 V

Vcc Undervoltage Hysteresis 0.35 V

Bias Current (Iin) FB = 1.3V 3.7 4.5 mA

Shutdown Current (Iin) SD = 0V 57 85 µA

SHUTDOWN THRESHOLDS

Shutdown Threshold (SD Increasing) 0.47 0.7 0.9 V

Shutdown Hysteresis 0.1 V

Standby Threshold (Standby Increasing) 1.17 1.225 1.28 V

Standby Hysteresis 0.1 V

SD Pull-up Current Source 5 µA

SWITCH CHARACTERSICS

Buck Switch Rds(on) 750 1500 mΩ

BOOST UVLO 4 V

BOOST UVLO Hysteresis 0.56 V

Pre-charge Switch Rds(on) 70 Ω

Pre-charge Switch on-time 250 ns

CURRENT LIMIT

Cycle by Cycle Current Limit RAMP = 0V 0.6 0.7 0.8 A

Cycle by Cycle Current Limit Delay RAMP = 2.5V 75 ns

SOFT-START

SS Current Source 710 14 µA

OSCILLATOR

Frequency1 180 200 220 kHz

Frequency2 RT = 11kΩ425 485 545 kHz

SYNC Source Impedance 11 kΩ

SYNC Sink Impedance 110 Ω

SYNC Threshold (falling) 1.3 V

SYNC Frequency RT = 11kΩ550 kHz

SYNC Pulse Width Minimum 15 ns

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LM5574/LM5574Q

Symbol Parameter Conditions Min Typ Max Units

RAMP GENERATOR

Ramp Current 1 Vin = 60V, Vout=10V 467 550 633 µA

Ramp Current 2 Vin = 10V, Vout=10V 36 50 64 µA

PWM COMPARATOR

Forced Off-time 416 500 575 ns

Min On-time 80 ns

COMP to PWM Comparator Offset 0.7 V

ERROR AMPLIFIER

Feedback Voltage Vfb = COMP 1.207 1.225 1.243 V

FB Bias Current 17 nA

DC Gain 70 dB

COMP Sink / Source Current 3 mA

Unity Gain Bandwidth 3 MHz

DIODE SENSE RESISTANCE

DSENSE 250 mΩ

THERMAL SHUTDOWN

Tsd Thermal Shutdown Threshold 165 °C

Thermal Shutdown Hysteresis 25 °C

THERMAL RESISTANCE

θJC Junction to Case 30 °C/W

θJA Junction to Ambient 90 °C/W

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the

device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin.

Note 3: Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlation using Statistical

Quality Control (SQC) methods. Limits are used to calculate National’s Average Outgoing Quality Level (AOQL).

Typical Performance Characteristics

Oscillator Frequency vs RT

20212020

Oscillator Frequency vs Temperature

FOSC = 200kHz

20212021

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LM5574/LM5574Q

Soft Start Current vs Temperature

20212022

VCC vs ICC

VIN = 12V

20212023

VCC vs VIN

RL = 7kΩ

20212024

Error Amplifier Gain/Phase

AVCL = 101

20212025

Demoboard Efficiency vs IOUT and VIN

20212026

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LM5574/LM5574Q

Typical Application Circuit and Block Diagram

20212003

FIGURE 1.

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LM5574/LM5574Q

Detailed Operating Description

The LM5574 switching regulator features all of the functions

necessary to implement an efficient high voltage buck regu-

lator using a minimum of external components. This easy to

use regulator integrates a 75V N-Channel buck switch with

an output current capability of 0.5 Amps. The regulator control

method is based on current mode control utilizing an emulat-

ed current ramp. Peak current mode control provides inherent

line voltage feed-forward, cycle-by-cycle current limiting, and

ease of loop compensation. The use of an emulated control

ramp reduces noise sensitivity of the pulse-width modulation

circuit, allowing reliable processing of very small duty cycles

necessary in high input voltage applications. The operating

frequency is user programmable from 50kHz to 500kHz. An

oscillator synchronization pin allows multiple LM5574 regula-

tors to self synchronize or be synchronized to an external

clock. The output voltage can be set as low as 1.225V. Fault

protection features include, current limiting, thermal shutdown

and remote shutdown capability. The device is available in the

TSSOP-16 package.

The functional block diagram and typical application of the

LM5574 are shown in Figure 1. The LM5574 can be applied

in numerous applications to efficiently step-down a high, un-

regulated input voltage. The device is well suited for telecom,

industrial and automotive power bus voltage ranges.

High Voltage Start-Up Regulator

The LM5574 contains a dual-mode internal high voltage start-

up regulator that provides the Vcc bias supply for the PWM

controller and boot-strap MOSFET gate driver. The input pin

(VIN) can be connected directly to the input voltage, as high

as 75 Volts. For input voltages below 9V, a low dropout switch

connects Vcc directly to Vin. In this supply range, Vcc is ap-

proximately equal to Vin. For Vin voltage greater than 9V, the

low dropout switch is disabled and the Vcc regulator is en-

abled to maintain Vcc at approximately 7V. The wide operat-

ing range of 6V to 75V is achieved through the use of this dual

mode regulator.

The output of the Vcc regulator is current limited to 25mA.

Upon power up, the regulator sources current into the capac-

itor connected to the VCC pin. When the voltage at the VCC

pin exceeds the Vcc UVLO threshold of 5.35V and the SD pin

is greater than 1.225V, the output switch is enabled and a soft-

start sequence begins. The output switch remains enabled

until Vcc falls below 5.0V or the SD pin falls below 1.125V.

An auxiliary supply voltage can be applied to the Vcc pin to

reduce the IC power dissipation. If the auxiliary voltage is

greater than 7.3V, the internal regulator will essentially

shut off, reducing the IC power dissipation. The Vcc regulator

series pass transistor includes a diode between Vcc and Vin

that should not be forward biased in normal operation. There-

fore the auxiliary Vcc voltage should never exceed the Vin

voltage.

In high voltage applications extra care should be taken to en-

sure the VIN pin does not exceed the absolute maximum

voltage rating of 76V. During line or load transients, voltage

ringing on the Vin line that exceeds the Absolute Maximum

Ratings can damage the IC. Both careful PC board layout and

the use of quality bypass capacitors located close to the VIN

and GND pins are essential.

20212004

FIGURE 2. Vin and Vcc Sequencing

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LM5574/LM5574Q

Shutdown / Standby

The LM5574 contains a dual level Shutdown (SD) circuit.

When the SD pin voltage is below 0.7V, the regulator is in a

low current shutdown mode. When the SD pin voltage is

greater than 0.7V but less than 1.225V, the regulator is in

standby mode. In standby mode the Vcc regulator is active

but the output switch is disabled. When the SD pin voltage

exceeds 1.225V, the output switch is enabled and normal op-

eration begins. An internal 5µA pull-up current source config-

ures the regulator to be fully operational if the SD pin is left

open.

An external set-point voltage divider from VIN to GND can be

used to set the operational input range of the regulator. The

divider must be designed such that the voltage at the SD pin

will be greater than 1.225V when Vin is in the desired oper-

ating range. The internal 5µA pull-up current source must be

included in calculations of the external set-point divider. Hys-

teresis of 0.1V is included for both the shutdown and standby

thresholds. The SD pin is internally clamped with a 1kΩ re-

sistor and an 8V zener clamp. The voltage at the SD pin

should never exceed 14V. If the voltage at the SD pin exceeds

8V, the bias current will increase at a rate of 1 mA/V.

The SD pin can also be used to implement various remote

enable / disable functions. Pulling the SD pin below the 0.7V

threshold totally disables the controller. If the SD pin voltage

is above 1.225V the regulator will be operational.

Oscillator and Sync Capability

The LM5574 oscillator frequency is set by a single external

resistor connected between the RT pin and the AGND pin.

The RT resistor should be located very close to the device and

connected directly to the pins of the IC (RT and AGND).To

set a desired oscillator frequency (F), the necessary value for

the RT resistor can be calculated from the following equation:

The SYNC pin can be used to synchronize the internal oscil-

lator to an external clock. The external clock must be of

higher frequency than the free-running frequency set by the

RT resistor. A clock circuit with an open drain output is the

recommended interface from the external clock to the SYNC

pin. The clock pulse duration should be greater than 15ns.

20212005

FIGURE 3. Sync from External Clock

20212006

FIGURE 4. Sync from Multiple Devices

Multiple LM5574 devices can be synchronized together sim-

ply by connecting the SYNC pins together. In this configura-

tion all of the devices will be synchronized to the highest

frequency device. The diagram in Figure 5 illustrates the

SYNC input/output features of the LM5574. The internal os-

cillator circuit drives the SYNC pin with a strong pull-down /

weak pull-up inverter. When the SYNC pin is pulled low either

by the internal oscillator or an external clock, the ramp cycle

of the oscillator is terminated and a new oscillator cycle be-

gins. Thus, if the SYNC pins of several LM5574 IC’s are

connected together, the IC with the highest internal clock fre-

quency will pull the connected SYNC pins low first and termi-

nate the oscillator ramp cycles of the other IC’s. The LM5574

with the highest programmed clock frequency will serve as

the master and control the switching frequency of the all the

devices with lower oscillator frequency.

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LM5574/LM5574Q

20212007

FIGURE 5. Simplified Oscillator Block Diagram and SYNC I/O Circuit

Error Amplifier and PWM

Comparator

The internal high gain error amplifier generates an error signal

proportional to the difference between the regulated output

voltage and an internal precision reference (1.225V). The

output of the error amplifier is connected to the COMP pin

allowing the user to provide loop compensation components,

generally a type II network, as illustrated in Figure 1. This

network creates a pole at DC, a zero and a noise reducing

high frequency pole. The PWM comparator compares the

emulated current sense signal from the RAMP generator to

the error amplifier output voltage at the COMP pin.

RAMP Generator

The ramp signal used in the pulse width modulator for current

mode control is typically derived directly from the buck switch

current. This switch current corresponds to the positive slope

portion of the output inductor current. Using this signal for the

PWM ramp simplifies the control loop transfer function to a

single pole response and provides inherent input voltage

feed-forward compensation. The disadvantage of using the

buck switch current signal for PWM control is the large leading

edge spike due to circuit parasitics that must be filtered or

blanked. Also, the current measurement may introduce sig-

nificant propagation delays. The filtering, blanking time and

propagation delay limit the minimum achievable pulsewidth.

In applications where the input voltage may be relatively large

in comparison to the output voltage, controlling small

pulsewidths and duty cycles is necessary for regulation. The

LM5574 utilizes a unique ramp generator, which does not ac-

tually measure the buck switch current but rather reconstructs

the signal. Reconstructing or emulating the inductor current

provides a ramp signal to the PWM comparator that is free of

leading edge spikes and measurement or filtering delays. The

current reconstruction is comprised of two elements; a sam-

ple & hold DC level and an emulated current ramp.

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LM5574/LM5574Q

20212008

FIGURE 6. Composition of Current Sense Signal

The sample & hold DC level illustrated in Figure 6 is derived

from a measurement of the re-circulating Schottky diode an-

ode current. The re-circulating diode anode should be con-

nected to the IS pin. The diode current flows through an

internal current sense resistor between the IS and PGND

pins. The voltage level across the sense resistor is sampled

and held just prior to the onset of the next conduction interval

of the buck switch. The diode current sensing and sample &

hold provide the DC level of the reconstructed current signal.

The positive slope inductor current ramp is emulated by an

external capacitor connected from the RAMP pin to AGND

and an internal voltage controlled current source. The ramp

current source that emulates the inductor current is a function

of the Vin and Vout voltages per the following equation:

IRAMP = (10µ x (Vin – Vout)) + 50µA

Proper selection of the RAMP capacitor depends upon the

selected value of the output inductor. The value of CRAMP can

be selected from: CRAMP = L x 5 x 10-6, where L is the value

of the output inductor in Henrys. With this value, the scale

factor of the emulated current ramp will be approximately

equal to the scale factor of the DC level sample and hold

(2.0V / A). The CRAMP capacitor should be located very close

to the device and connected directly to the pins of the IC

(RAMP and AGND).

For duty cycles greater than 50%, peak current mode control

circuits are subject to sub-harmonic oscillation. Sub-harmonic

oscillation is normally characterized by observing alternating

wide and narrow pulses at the switch node. Adding a fixed

slope voltage ramp (slope compensation) to the current sense

signal prevents this oscillation. The 50µA of offset current

provided from the emulated current source adds some fixed

slope to the ramp signal. In some high output voltage, high

duty cycle applications, additional slope may be required. In

these applications, a pull-up resistor may be added between

the VCC and RAMP pins to increase the ramp slope compen-

sation.

For VOUT > 7.5V:

Calculate optimal slope current, IOS = VOUT x 10µA/V.

For example, at VOUT = 10V, IOS = 100µA.

Install a resistor from the RAMP pin to VCC:

RRAMP = VCC / (IOS - 50µA)

20212045

FIGURE 7. RRAMP to VCC for VOUT > 7.5V

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LM5574/LM5574Q

Maximum Duty Cycle / Input Drop-

out Voltage

There is a forced off-time of 500ns implemented each cycle

to guarantee sufficient time for the diode current to be sam-

pled. This forced off-time limits the maximum duty cycle of the

buck switch. The maximum duty cycle will vary with the op-

erating frequency.

DMAX = 1 - Fs x 500ns

Where Fs is the oscillator frequency. Limiting the maximum

duty cycle will raise the input dropout voltage. The input

dropout voltage is the lowest input voltage required to main-

tain regulation of the output voltage. An approximation of the

input dropout voltage is:

Where VD is the voltage drop across the re-circulatory diode.

Operating at high switching frequency raises the minimum in-

put voltage necessary to maintain regulation.

Current Limit

The LM5574 contains a unique current monitoring scheme for

control and over-current protection. When set correctly, the

emulated current sense signal provides a signal which is pro-

portional to the buck switch current with a scale factor of 2.0

V / A. The emulated ramp signal is applied to the current limit

comparator. If the emulated ramp signal exceeds 1.4V (0.7A)

the present current cycle is terminated (cycle-by-cycle current

limiting). In applications with small output inductance and high

input voltage the switch current may overshoot due to the

propagation delay of the current limit comparator. If an over-

shoot should occur, the diode current sampling circuit will

detect the excess inductor current during the off-time of the

buck switch. If the sample & hold DC level exceeds the 1.4V

current limit threshold, the buck switch will be disabled and

skip pulses until the diode current sampling circuit detects the

inductor current has decayed below the current limit thresh-

old. This approach prevents current runaway conditions due

to propagation delays or inductor saturation since the inductor

current is forced to decay following any current overshoot.

Soft-Start

The soft-start feature allows the regulator to gradually reach

the initial steady state operating point, thus reducing start-up

stresses and surges. The internal soft-start current source,

set to 10µA, gradually increases the voltage of an external

soft-start capacitor connected to the SS pin. The soft-start

capacitor voltage is connected to the reference input of the

error amplifier. Various sequencing and tracking schemes

can be implemented using external circuits that limit or clamp

the voltage level of the SS pin.

In the event a fault is detected (over-temperature, Vcc UVLO,

SD) the soft-start capacitor will be discharged. When the fault

condition is no longer present a new soft-start sequence will

commence.

Boost Pin

The LM5574 integrates an N-Channel buck switch and asso-

ciated floating high voltage level shift / gate driver. This gate

driver circuit works in conjunction with an internal diode and

an external bootstrap capacitor. A 0.022µF ceramic capacitor,

connected with short traces between the BST pin and SW pin,

is recommended. During the off-time of the buck switch, the

SW pin voltage is approximately -0.5V and the bootstrap ca-

pacitor is charged from Vcc through the internal bootstrap

diode. When operating with a high PWM duty cycle, the buck

switch will be forced off each cycle for 500ns to ensure that

the bootstrap capacitor is recharged.

Under very light load conditions or when the output voltage is

pre-charged, the SW voltage will not remain low during the

off-time of the buck switch. If the inductor current falls to zero

and the SW pin rises, the bootstrap capacitor will not receive

sufficient voltage to operate the buck switch gate driver. For

these applications, the PRE pin can be connected to the SW

pin to pre-charge the bootstrap capacitor. The internal pre-

charge MOSFET and diode connected between the PRE pin

and PGND turns on each cycle for 250ns just prior to the onset

of a new switching cycle. If the SW pin is at a normal negative

voltage level (continuous conduction mode), then no current

will flow through the pre-charge MOSFET/diode.

Thermal Protection

Internal Thermal Shutdown circuitry is provided to protect the

integrated circuit in the event the maximum junction temper-

ature is exceeded. When activated, typically at 165°C, the

controller is forced into a low power reset state, disabling the

output driver and the bias regulator. This feature is provided

to prevent catastrophic failures from accidental device over-

heating.

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LM5574/LM5574Q

Application Information

EXTERNAL COMPONENTS

The procedure for calculating the external components is il-

lustrated with the following design example. The Bill of Mate-

rials for this design is listed in Table 1. The circuit shown in

Figure 1 is configured for the following specifications:

•VOUT = 5V

•VIN = 7V to 75V

•Fs = 300kHz

•Minimum load current (for CCM) = 100mA

•Maximum load current = 0.5A

R3 (RT)

RT sets the oscillator switching frequency. Generally, higher

frequency applications are smaller but have higher losses.

Operation at 300kHz was selected for this example as a rea-

sonable compromise for both small size and high efficiency.

The value of RT for 300kHz switching frequency can be cal-

culated as follows:

The nearest standard value of 21kΩ was chosen for RT.

The inductor value is determined based on the operating fre-

quency, load current, ripple current, and the minimum and

maximum input voltage (VIN(min), VIN(max)).

20212010

FIGURE 8. Inductor Current Waveform

To keep the circuit in continuous conduction mode (CCM), the

maximum ripple current IRIPPLE should be less than twice the

minimum load current, or 0.2Ap-p. Using this value of ripple

current, the value of inductor (L1) is calculated using the fol-

lowing:

This procedure provides a guide to select the value of L1. The

nearest standard value (100µH) will be used. L1 must be rated

for the peak current (IPK+) to prevent saturation. During normal

loading conditions, the peak current occurs at maximum load

current plus maximum ripple. During an overload condition

the peak current is limited to 0.7A nominal (0.85A maximum).

The selected inductor (see Table 1) has a conservative 1.0

Amp saturation current rating. For this manufacturer, the sat-

uration rating is defined as the current necessary for the

inductance to reduce by 30%, at 20°C.

C3 (CRAMP)

With the inductor value selected, the value of C3 (CRAMP)

necessary for the emulation ramp circuit is:

CRAMP = L x 5 x 10-6

Where L is in Henrys

With L1 selected for 100µH the recommended value for C3 is

470pF (nearest standard value).

The output capacitor, C9 smoothes the inductor ripple current

and provides a source of charge for transient loading condi-

tions. For this design a 22µF ceramic capacitor was selected.

The ceramic capacitor provides ultra low ESR to reduce the

output ripple voltage and noise spikes. An approximation for

the output ripple voltage is:

A Schottky type re-circulating diode is required for all LM5574

applications. Ultra-fast diodes are not recommended and may

result in damage to the IC due to reverse recovery current

transients. The near ideal reverse recovery characteristics

and low forward voltage drop are particularly important diode

characteristics for high input voltage and low output voltage

applications common to the LM5574. The reverse recovery

characteristic determines how long the current surge lasts

each cycle when the buck switch is turned on. The reverse

recovery characteristics of Schottky diodes minimize the peak

instantaneous power in the buck switch occurring during turn-

on each cycle. The resulting switching losses of the buck

switch are significantly reduced when using a Schottky diode.

The reverse breakdown rating should be selected for the

maximum VIN, plus some safety margin.

The forward voltage drop has a significant impact on the con-

version efficiency, especially for applications with a low output

voltage. “Rated” current for diodes vary widely from various

manufacturers. The worst case is to assume a short circuit

load condition. In this case the diode will carry the output cur-

rent almost continuously. For the LM5574 this current can be

as high as 0.7A. Assuming a worst case 1V drop across the

diode, the maximum diode power dissipation can be as high

as 0.7W. For the reference design a 100V Schottky in a SMA

package was selected.

The regulator supply voltage has a large source impedance

at the switching frequency. Good quality input capacitors are

necessary to limit the ripple voltage at the VIN pin while sup-

plying most of the switch current during the on-time. When the

buck switch turns on, the current into the VIN pin steps to the

lower peak of the inductor current waveform, ramps up to the

peak value, then drops to zero at turn-off. The average current

into VIN during the on-time is the load current. The input ca-

pacitance should be selected for RMS current rating and

13 www.national.com

LM5574/LM5574Q

minimum ripple voltage. A good approximation for the re-

quired ripple current rating necessary is IRMS > IOUT / 2.

Quality ceramic capacitors with a low ESR should be selected

for the input filter. To allow for capacitor tolerances and volt-

age effects, one 1.0µF, 100V ceramic capacitor will be used.

If step input voltage transients are expected near the maxi-

mum rating of the LM5574, a careful evaluation of ringing and

possible spikes at the device VIN pin should be completed.

An additional damping network or input voltage clamp may be

required in these cases.

The capacitor at the VCC pin provides noise filtering and sta-

bility for the VCC regulator. The recommended value of C8

should be no smaller than 0.1µF, and should be a good qual-

ity, low ESR, ceramic capacitor. A value of 0.47µF was se-

lected for this design.

The bootstrap capacitor between the BST and the SW pins

supplies the gate current to charge the buck switch gate at

turn-on. The recommended value of C7 is 0.022µF, and

should be a good quality, low ESR, ceramic capacitor.

The capacitor at the SS pin determines the soft-start time, i.e.

the time for the reference voltage and the output voltage, to

reach the final regulated value. The time is determined from:

For this application, a C4 value of 0.01µF was chosen which

corresponds to a soft-start time of 1ms.

R5, R6

R5 and R6 set the output voltage level, the ratio of these re-

sistors is calculated from:

R5/R6 = (VOUT / 1.225V) - 1

For a 5V output, the R5/R6 ratio calculates to 3.082. The re-

sistors should be chosen from standard value resistors, a

good starting point is selection in the range of 1.0kΩ - 10kΩ.

Values of 5.11kΩ for R5, and 1.65kΩ for R6 were selected.

R1, R2, C2

A voltage divider can be connected to the SD pin to set a

minimum operating voltage Vin(min) for the regulator. If this

feature is required, the easiest approach to select the divider

resistor values is to select a value for R1 (between 10kΩ and

100kΩ recommended) then calculate R2 from:

Capacitor C2 provides filtering for the divider. The voltage at

the SD pin should never exceed 8V, when using an external

set-point divider it may be necessary to clamp the SD pin at

high input voltage conditions. The reference design utilizes

the full range of the LM5574 (6V to 75V); therefore these

components can be omitted. With the SD pin open circuit the

LM5574 responds once the Vcc UVLO threshold is satisfied.

R4, C5, C6

These components configure the error amplifier gain charac-

teristics to accomplish a stable overall loop gain. One advan-

tage of current mode control is the ability to close the loop with

only two feedback components, R4 and C5. The overall loop

gain is the product of the modulator gain and the error ampli-

fier gain. The DC modulator gain of the LM5574 is as follows:

DC Gain(MOD) = Gm(MOD) x RLOAD = 0.5 x RLOAD

The dominant low frequency pole of the modulator is deter-

mined by the load resistance (RLOAD,) and output capacitance

(COUT). The corner frequency of this pole is:

fp(MOD) = 1 / (2π RLOAD COUT)

For RLOAD = 20Ω and COUT = 22µF then fp(MOD) = 362Hz

DC Gain(MOD) = 0.5 x 20 = 20dB

For the design example of Figure 1 the following modulator

gain vs. frequency characteristic was measured as shown in

Figure 9.

20212015

FIGURE 9. Gain and Phase of Modulator

RLOAD = 20 Ohms and COUT = 22µF

Components R4 and C5 configure the error amplifier as a type

II configuration which has a pole at DC and a zero at fZ = 1 /

(2πR4C5). The error amplifier zero cancels the modulator

pole leaving a single pole response at the crossover frequen-

cy of the loop gain. A single pole response at the crossover

frequency yields a very stable loop with 90 degrees of phase

margin.

For the design example, a target loop bandwidth (crossover

frequency) of 25kHz was selected. The compensation net-

work zero (fZ) should be selected at least an order of magni-

tude less than the target crossover frequency. This constrains

the product of R4 and C5 for a desired compensation network

zero 1 / (2π R4 C5) to be less than 2kHz. Increasing R4, while

proportionally decreasing C5, increases the error amp gain.

Conversely, decreasing R4 while proportionally increasing

C5, decreases the error amp gain. For the design example

C5 was selected for 0.022µF and R4 was selected for

24.9kΩ. These values configure the compensation network

www.national.com 14

LM5574/LM5574Q

zero at 290Hz. The error amp gain at frequencies greater than

fZ is: R4 / R5, which is approximately 5 (14dB).

20212016

FIGURE 10. Error Amplifier Gain and Phase

The overall loop can be predicted as the sum (in dB) of the

modulator gain and the error amp gain.

20212017

FIGURE 11. Overall Loop Gain and Phase

If a network analyzer is available, the modulator gain can be

measured and the error amplifier gain can be configured for

the desired loop transfer function. If a network analyzer is not

available, the error amplifier compensation components can

be designed with the guidelines given. Step load transient

tests can be performed to verify acceptable performance. The

step load goal is minimum overshoot with a damped re-

sponse. C6 can be added to the compensation network to

decrease noise susceptibility of the error amplifier. The value

of C6 must be sufficiently small since the addition of this ca-

pacitor adds a pole in the error amplifier transfer function. This

pole must be well beyond the loop crossover frequency. A

good approximation of the location of the pole added by C6

is: fp2 = fz x C5 / C6.

BIAS POWER DISSIPATION REDUCTION

Buck regulators operating with high input voltage can dissi-

pate an appreciable amount of power for the bias of the IC.

The VCC regulator must step-down the input voltage VIN to a

nominal VCC level of 7V. The large voltage drop across the

VCC regulator translates into a large power dissipation within

the Vcc regulator. There are several techniques that can sig-

nificantly reduce this bias regulator power dissipation. Figure

12 and Figure 13 depict two methods to bias the IC from the

output voltage. In each case the internal Vcc regulator is used

to initially bias the VCC pin. After the output voltage is estab-

lished, the VCC pin potential is raised above the nominal 7V

regulation level, which effectively disables the internal VCC

regulator. The voltage applied to the VCC pin should never

exceed 14V. The VCC voltage should never be larger than the

VIN voltage.

15 www.national.com

LM5574/LM5574Q

20212018

FIGURE 12. VCC Bias from VOUT for 8V < VOUT < 14V

20212019

FIGURE 13. VCC Bias with Additional Winding on the Output Inductor

www.national.com 16

LM5574/LM5574Q

PCB LAYOUT AND THERMAL CONSIDERATIONS

The circuit in Figure 1 serves as both a block diagram of the

LM5574 and a typical application board schematic for the

LM5574. In a buck regulator there are two loops where cur-

rents are switched very fast. The first loop starts from the input

capacitors, to the regulator VIN pin, to the regulator SW pin,

to the inductor then out to the load. The second loop starts

from the output capacitor ground, to the regulator PGND pins,

to the regulator IS pins, to the diode anode, to the inductor

and then out to the load. Minimizing the loop area of these

two loops reduces the stray inductance and minimizes noise

and possible erratic operation. A ground plane in the PC

board is recommended as a means to connect the input filter

capacitors to the output filter capacitors and the PGND pins

of the regulator. Connect all of the low power ground connec-

tions (CSS, RT, CRAMP) directly to the regulator AGND pin.

Connect the AGND and PGND pins together through the top-

side copper trace. Place several vias in this trace to the

ground plane.

The two highest power dissipating components are the re-

circulating diode and the LM5574 regulator IC. The easiest

method to determine the power dissipated within the LM5574

is to measure the total conversion losses (Pin – Pout) then

subtract the power losses in the Schottky diode, output in-

ductor and snubber resistor. An approximation for the Schot-

tky diode loss is P = (1-D) x Iout x Vfwd. An approximation for

the output inductor power is P = IOUT2 x R x 1.1, where R is

the DC resistance of the inductor and the 1.1 factor is an ap-

proximation for the AC losses. If a snubber is used, an ap-

proximation for the damping resistor power dissipation is P =

Vin2 x Fsw x Csnub, where Fsw is the switching frequency

and Csnub is the snubber capacitor.

The most significant variables that affect the power dissipated

by the LM5574 are the output current, input voltage and op-

erating frequency. The power dissipated while operating near

the maximum output current and maximum input volatge can

be appreciable. The operating frequency of the LM5574 eval-

uation board has been designed for 300kHz. When operating

at 0.5A output current with a 70V input the power dissipation

of the LM5574 regulator is approximately 0.6W.

The junction-to-ambient thermal resistance of the LM5574 will

vary with the application. The most significant variables are

the area of copper in the PC board, and the amount of forced

air cooling provided. The junction-to-ambient thermal resis-

tance of the LM5574 mounted in the evaluation board varies

from 90°C/W with no airflow to 60°C/W with 900 LFM (Linear

Feet per Minute). With a 25°C ambient temperature and no

airflow, the predicted junction temperature for the LM5574 will

be 25 + ((90 x 0.6) = 79°C. If the evaluation board is operated

at 0.5A output current, 70V input voltage and high ambient

temperature for a prolonged period of time the thermal shut-

down protection within the IC may activate. The IC will turn

off allowing the junction to cool, followed by restart with the

soft-start capacitor reset to zero.

17 www.national.com

LM5574/LM5574Q

TABLE 1. 5V, 0.5A Demo Board Bill of Materials

Item Part Number Description Value

C 1 C3225X7R2A105M CAPACITOR, CER, TDK 1µ, 100V

C 2 OPEN NOT USED

C 3 C0805A471K1GAC CAPACITOR, CER, KEMET 470p, 100V

C 4 C2012X7R2A103K CAPACITOR, CER, TDK 0.01µ, 100V

C 5 C2012X7R2A223K CAPACITOR, CER, TDK 0.022µ, 100V

C 6 OPEN NOT USED

C 7 C2012X7R2A223K CAPACITOR, CER, TDK 0.022µ, 100V

C 8 C2012X7R1C474M CAPACITOR, CER, TDK 0.47µ, 16V

C 9 C3225X7R1C226M CAPACITOR, CER, TDK 22µ, 16V

D 1 CMSH2-100M DIODE, 100V, CENTRAL

L 1 DR74-101 INDUCTOR, COOPER 100µH

R 1 OPEN NOT USED

R 2 OPEN NOT USED

R 3 CRCW08052102F RESISTOR 21kΩ

R 4 CRCW08052492F RESISTOR 24.9kΩ

R 5 CRCW08055111F RESISTOR 5.11kΩ

R 6 CRCW08051651F RESISTOR 1.65kΩ

U 1 LM5574 REGULATOR, NATIONAL SEMICONDUCTOR

www.national.com 18

LM5574/LM5574Q

PCB Layout

20212029

Component Side

20212030

Solder Side

20212031

Silkscreen

19 www.national.com

LM5574/LM5574Q

Physical Dimensions inches (millimeters) unless otherwise noted

16-Lead TSSOP Package

NS Package Number MTC16

www.national.com 20

LM5574/LM5574Q

Notes

21 www.national.com

LM5574/LM5574Q

Notes

LM5574/LM5574Q SIMPLE SWITCHER® 75V, 0.5A Step-Down Switching Regulator

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